Priority messaging method for a discrete multitone spread spectrum communications system

ABSTRACT

A discrete multitone stacked-carrier spread spectrum communication method is based on frequency domain spreading including multiplication of a baseband signal by a set of superimposed, or stacked, complex sinusoid carrier waves. In a preferred embodiment, the spreading involves energizing the bins of a large Fast Fourier transform (FFT). This provides a considerable savings in computational complexity for moderate output FFT sizes. Point-to-multipoint and multipoint-to-multipoint (nodeless) network topologies are possible. A code-nulling method is included for interference cancellation and enhanced signal separation by exploiting the spectral diversity of the various sources. The basic method may be extended to include multielement antenna array nulling methods for interference cancellation and enhanced signal separation using spatial separation. Such methods permit directive and retrodirective transmission systems that adapt or can be adapted to the radio environment. Such systems are compatible with bandwidth-on-demand and higher-order modulation formats and use advanced adaptation algorithms. In a specific embodiment the spectral and spatial components of the adaptive weights are calculated in a unified operation based on the mathematical analogy between the spectral and spatial descriptions of the airlink.

This application is a division of U.S. patent application Ser. No.08/993,721, filed Dec. 18, 1997 now U.S. Pat. No. 6,359,923.

FIELD OF THE INVENTION

This invention involves communications methods that make very efficientuse of available spectral bandwidth by a combination of multiple accesstechniques.

BACKGROUND OF THE INVENTION

Communication systems that operate over limited spectral bandwidths mustmake highly efficient use of the scarce bandwidth resource to provideacceptable service to a large population of users. Examples of suchcommunications systems that deal with high user demand and scarcebandwidth resources are wireless communications systems, such ascellular and personal communications systems.

Various techniques have been suggested for use in such systems toincrease bandwidth-efficiency—the amount of information that can beeffectively transmitted within a given spectral bandwidth. Many of thesetechniques involve reusing the same communication resources for multipleusers while maintaining the identity of each user's signal. Thesetechniques are generically referred to as multiple access techniques orprotocols. Among these multiple access protocols are Time DivisionMultiple Access (TDMA), Code Division Multiple Access (CDMA), SpaceDivision Multiple Access (SDMA), and Frequency Division Multiple Access(FDMA). The technical foundations of these multiple access protocols arediscussed, for example, in the recent book by Rappaport entitled“Wireless Communications Principles and Practice”, Prentice Hall, 1996.

The Time Division Multiple Access (TDMA) protocol involves thetransmission of information from a multiplicity of users on one assignedfrequency bandwidth by time division multiplexing the information fromthe various users. In this multiplexing scheme, particular time slotsare devoted to specific users. Knowledge of the time slot during whichany specific information is transmitted, permits the separation andreconstruction of each user's message at the receiving end of thecommunication channel.

The Code Division Multiple Access (CDMA) protocol involves the use of aunique code to distinguish each user's data signal from other users'data signals. Knowledge of the unique code with which any specificinformation is transmitted, permits the separation and reconstruction ofeach user's message at the receiving end of the communication channel.There are four types of CDMA protocols, classified by the specifictechnique that is used to spread the user's data over a wide portion ofthe frequency spectrum: direct sequence (or pseudo-noise), frequencyhopping, time hopping, and hybrid systems. The technical foundations forCDMA protocols are discussed, for example, in the recent book by Prasadentitled “CDMA for Wireless Personal Communications”, Artech House,1996.

The Direct Sequence CDMA (DS-CDMA) protocol involves the spreading of auser's data signal over a wide portion of the frequency spectrum bymodulating the data signal with a unique code signal that is of higherbandwidth than the data signal. The frequency of the code signal ischosen to be much larger than the frequency of the data signal. The datasignal is directly modulated by the code signal and the resultingencoded data signal modulates a single, wideband carrier thatcontinuously covers a wide frequency range. After transmission of theDS-CDMA modulated carrier signal, the receiver uses a locally generatedversion of the user's unique code signal to demodulate the receivedsignal and obtain a reconstructed data signal. The receiver is thus ableto extract the user's data signal from a modulated carrier that bearsmany other users' data signals.

The Frequency Hopping Spread Spectrum (FHSS) protocol involves the useof a unique code to change the value of a narrowband carrier frequencyfor successive bursts of the user's data signal. The value of thecarrier frequency varies in time over a wide range of the frequencyspectrum in accordance with the unique code. CDMA protocols are closelyrelated to spread spectrum technology and the term Spread SpectrumMultiple Access (SSMA) is also used for CDMA protocols such as DS-CDMAand FHSS that use a relatively wide frequency range over which todistribute a relatively narrowband data signal.

The Time Hopping CDMA (TH-CDMA) protocol involves the use of a single,narrow bandwidth, carrier frequency to send bursts of the user's dataduring intervals determined by the user's unique code.

Hybrid CDMA systems employ a combination of two or more CDMA protocols,such as direct sequence/frequency hopping (DS/FH), direct sequence/timehopping (DS/TH), frequency hopping/time hopping (FH/TH), and directsequence/frequency hopping/time hopping (DS/FH/TH).

The CDMA protocols modulate each user's information with a differentcode unique to that user. Each user's information is separated andreconstructed at the receiving end of the communication channel byisolating that portion of the multiplexed signal that correlates withthe user's code. In specific embodiments, orthogonal codes are used,permitting the complete separation of information associated withdifferent codes, without cross-talk. If orthogonal codes are notemployed, “code nulling” may be employed to limit interference due tocorrelation between various codes. This technique involves the judiciousselection of codes that, though non-orthogonal, result in only minimalcross-talk.

The Space Division Multiple Access (SDMA) transmission protocol involvesthe formation of directed beams of energy, whose radiation patterns donot overlap spatially with each other, to communicate with users atdifferent locations. Adaptive antenna arrays can be driven in phasedpatterns to simultaneously steer energy in the direction of selectedreceivers. With such a transmission technique, the other multiplexingschemes can be reused in each of the separately directed beams. Forexample, the same specific CDMA codes can be used in two differentspatially separated beams. Accordingly, if the beams do not overlap eachother, different users can be assigned the same code as long as they canbe uniquely identified by a specific beam/code combination.

The SDMA receive protocol involves the use of multi-element adaptiveantenna arrays to direct the receiving sensitivity of the array towardselected transmitting sources. Digital beamforming is used to processthe signals received by the adaptive antenna array and to separateinterference and noise from genuine signals received from any givendirection. For a receiving station, received RF signals at each antennaelement in the array are sampled and digitized. The digital basebandsignals then represent the amplitudes and phases of the RF signalsreceived at each antenna element in the array. Digital signal processingtechniques are then applied to the digital stream from each antennaelement in the array. The process of beamforming involves theapplication of weight values to the digital signals from each antennaelement, thereby adjusting the numerical representation of theiramplitudes and phases such that when added together, they form thedesired beam—i.e., the desired directional receive sensitivity. The beamthus formed is a digital representation within the computer of thephysical RF signals received by the antenna array from any givendirection. The process of null steering at the transmitter is used toposition the spatial direction of null regions in the pattern of thetransmitted RF energy. The process of null steering at the receiver is adigital signal processing technique to control the effective directionof nulls in the receiver's gain or sensitivity. Both processes areintended to minimize inter-beam spatial interference. SDMA techniquesusing multi-element antenna arrays to form directed beams are disclosedin the context of mobile communications in Swales et. al., IEEE Trans.Veh. Technol. Vol. 39. No. 1 February 1990, and in U.S. Pat. No.5,515,378. The technical foundations for SDMA protocols using adaptiveantenna arrays are discussed, for example, in the recent book by Litvaand Lo entitled “Digital Beamforming in Wireless Communications”, ArtechHouse, 1996.

The Frequency Division Multiple Access (FDMA) protocol services amultiplicity of users over one frequency band by devoting particularfrequency slots to specific users, i.e., by frequency divisionmultiplexing the information associated with different users. Knowledgeof the frequency slot in which any specific information resides permitsreconstruction of each user's information at the receiving end of thecommunication channel.

Orthogonal Frequency Division Multiplexing (OFDM) addresses a problemthat is faced, for example, when pulsed signals are transmitted in anFDMA format. In accordance with principles well known in thecommunication sciences, the limited time duration of such signalsinherently broadens the bandwidth of the signal in frequency space.Accordingly, different frequency channels may significantly overlap,defeating the use of frequency as a user-identifying-parameter, theprinciple upon which FDMA is based. However, as discussed immediatelybelow, pulsed information that is transmitted on specific frequenciescan be separated, in accordance with OFDM principles, despite the factthat the frequency channels overlap due to the limited time duration ofthe signals.

OFDM requires a specific relationship between the data rate and thecarrier frequencies. Specifically, the total signal frequency band isdivided into N frequency sub-channels, each of which has the same datarate 1/T. These data streams are then multiplexed onto a multiplicity ofcarriers that are separated in frequency by 1/T. Multiplexing signalsunder these constraints results in each carrier having a frequencyresponse that has zeroes at multiples of 1/T. Therefore, there is nointerference between the various carrier channels, despite the fact thatthe channels overlap each other because of the broadening associatedwith the data rate. OFDM is disclosed, for example, by Chang in BellSys. Tech. Jour., Vol. 45, pp. 1775-1796, December 1966, and in U.S.Pat. No. 4,488,445.

Parallel Data Transmission is a technique related to FDMA. It is alsoreferred to as Multitone Transmission (MT), Discrete MultitoneTransmission (DMT) or Multi-Carrier Transmission (MCT). Parallel DataTransmission has significant calculational advantages over simple FDMA.In this technique, each user's information is divided and transmittedover different frequencies, or “tones”, rather than over a singlefrequency, as in standard FDMA. In an example of this technique, inputdata at NF bits per second are grouped into blocks of N bits at a datarate of F bits per second. N carriers or “tones” are then used totransmit these bits, each carrier transmitting F bits per second. Thecarriers can be spaced in accordance with the principles of OFDM.

A benefit of parallel data transmission derives from certaincomputational advantages associated with this transmission technique.Specifically, it can be shown that a parallel data signal is equivalentto the Fourier transform of the original serial data train and that thedemodulation of the tones is equivalent to the inverse Fouriertransform. This has led to the advantageous use of fast Fouriertransform techniques (FFT) in implementing this technique, rather thanthe use of an expensive system of sinusoidal generators, modulators andcoherent demodulators. See, for example, Weinstein and Ebert, IEEETrans. on Comm. Tech., Vol. com-19, No. 5, October 1971, page 628.

Parallel data transmission can be used to service a multitude of usersby dedicating specific tones to specific users. In this technique,specific information can be uniquely associated with any particular userby transmitting information only on that user's assigned set offrequencies or tone set. The use of multiple frequencies for one userpermits the spreading of the signal over a wide, though discrete,portion of the frequency domain with the benefits familiar from spreadspectrum communications. See U.S. Pat. No. 5,410,538 issued to Roche andWyner.

Further multiplexing can be obtained by reusing the same set offrequencies or tone set for different users by modulating the tone setbased on a user specific spreading code. Users assigned to the same toneset can then be distinguished by separating that portion of themultiplexed signals that correlate with their assigned code. See Yee,Linnartz, and Fettweis, “Multicarrier CDMA in indoor wireless radionetworks,” Proc. PIMRC '93, Yokohama, Japan, pp. 109-113, September1993.

Both the phase and the amplitude of the carrier can be varied torepresent the signal in multitone transmission. Accordingly, multitonetransmission can be implemented with M-ary digital modulation schemes.In an M-ary modulation scheme, two or more bits are grouped together toform symbols and one of the M possible signals is transmitted duringeach symbol period. Examples of M-ary digital modulation schemes includePhase Shift Keying (PSK), Frequency Shift Keying (FSK), and higher orderQuadrature Amplitude Modulation (QAM). In QAM a signal is represented bythe phase and amplitude of a carrier wave. In high order QAM, amultitude of points can be distinguished on a amplitude/phase plot. Forexample, in 64-ary QAM, 64 such points can be distinguished. Since sixbits of zeros and ones can take on 64 different combinations, a six-bitsequence of data symbols can, for example, be modulated onto a carrierin 64-ary QAM by transmitting only one value set of phase and amplitude,out of the possible 64 such sets.

Suggestions have been made to combine some of the above temporal andspectral multiplexing techniques. For example, in U.S. Pat. No.5,260,967, issued to Schilling, there is disclosed the combination ofTDMA and CDMA. In U.S. Pat. No. 5,291,475, issued to Bruckert, and inU.S. Pat. No. 5,319,634 issued to Bartholomew, the combination of TDMA,FDMA, and CDMA is suggested.

Other suggestions have been made to combine various temporal andspectral multiple-access techniques with spatial multiple-accesstechniques. For example, in U.S. Pat. No. 5,515,378, filed Dec. 12,1991, Roy suggests “separating multiple messages in the same frequency,code, or time channel using the fact that they are in different spatialchannels.” Roy suggests specific application of his technique to mobilecellular communications using an “antenna array”. Similar suggestionswere made by Swales et. al., in the IEEE Trans. Veh. Technol. Vol. 39.No. 1 February 1990, and by Davies et. al. in A.T.R., Vol. 22, No. 1,1988 and in Telecom Australia, Rev. Activities, 1985/1986 pp. 41-43.

In U.S. Pat. No. 5,260,968, filed Jun. 23, 1992, Gardner and Schellsuggest the use of communications channels that are “spectrallydisjoint” in conjunction with “spatially separable” radiation patterns.The radiation patterns are determined by restoring “self coherence”properties of the signal using an adaptive antenna array. “[A]n adaptiveantenna array at a base station is used in conjunction with signalprocessing through self coherence restoral to separate the temporallyand spectrally overlapping signals of users that arrive from differentspecific locations.” See the Abstract of the Invention. In this patent,however, adaptive analysis and self coherence restoral is only used todetermine the optimal beam pattern; “ . . . conventional spectralfilters . . . [are used] . . . to separate spatially inseparablefilters.”

Winters suggests “adaptive array processing” in which “[t]he frequencydomain data from a plurality of antennas are . . . combined for channelseparation and conversion to the time domain for demodulation.” See U.S.Pat. No. 5,481,570, filed Oct. 20, 1993, Column 1 lines 66-67 and Column2, lines 14-16.

Agee has shown that “the use of an M-element multiport antenna array atthe base station of any communication network can increase the frequencyreuse of the network by a factor of M and greatly broaden the range ofinput SINRs required for adequate demodulation . . . ” (“WirelessPersonal Communications: Trends and Challenges”, Rappaport, Woerner andReed, editors, Kluwer Academic Publishers, 1994, pp. 69-80, at page 69.See also, Proc. Virginia Tech. Third Symposium on Wireless PersonalCommunications, June 1993, pp. 15-1 to 15-12.) Agee asserts that in thisaspect of his work “[s]patial diversity can be exploited for anynetworking approach and modulation format, by employing a multiportadaptive antenna array to separate the time-coincident subscribersignals prior to the demodulation operation [underlining added].” op.cit. page 72. In that same work, Agee separately demonstrates that theproblem of receiving “signals over greatly disparate propagation ranges”. . . “can be overcome by exploiting the . . . spectral diversityinherent to the modulation format employed by typical communicationnetworks.” op. cit. page 69. Considering CDMA networks, Agee shows that“the single-antenna received data signal . . . can be transformed to . .. a vector sequence . . . [that] . . . bears a strong resemblance to thesignal generated by a narrowband antenna array receiving . . . spatiallycoherent signals in the presence of background interference.” op. cit.p. 76. The discussion is in terms of “CDMA networks employing an M-chipmodulation-on-symbol (MOS) DSSS spreading format . . . ” op. cit. p. 69.(DSSS is the abbreviation for the direct sequence spectrum spreading orDS-CDMA protocol.)

Gardner and Schell, in U.S. Pat. No. 5,260,968, filed Jun. 23, 1992,also suggest “time division multiplexing of the signal from the basestation and the users” . . . “[i]n order to use the same frequency forduplex communications . . . ” “[R]eception at the base station from allmobile units is temporally separated from transmission from the basestation to all mobile units.” Column 5, lines 44ff. In a similar vein,in U.S. Pat. No. 4,383,332 there is disclosed a wireless multi-elementadaptive antenna array SDMA system where all the required adaptivesignal processing is performed at baseband at the base station throughthe use of “time division retransmission techniques.”

Fazel, “Narrow-Band Interference Rejection in Orthogonal Multi-CarrierSpread-Spectrum Communications”, Record, 1994 Third Annual InternationalConference on Universal Personal Communications, IEEE, 1994, pp. 46-50describes a transmission scheme based on combined spread spectrum andOFDM. A plurality of subcarrier frequencies have components of thespreaded vector assigned to them to provide frequency-diversity at thereceiver site. The scheme uses frequency domain analysis to estimateinterference, which is used for weighting each received subcarrierbefore despreading. This results in switching off those subcarrierscontaining the interference.

Other disclosures of interest in this area include:

N. Yee, Jean-Paul M. G. Linnarta, G. Fettweis, “Multi-Carrier CDMA inIndoor Wireless Radio Networks”, IEICE Transactions on Communications,Vol. E77-B, No. 7 pp. 900-904, July 1994;

L. Vandendorpe, “Multitone Spread Spectrum Multiple AccessCommunications System in a Multipath Rician Fading Channel”, IEEETransactions on Vehicular Technology, Vol. 44 No.2, pp.327-337, May1995;

L. Vandendorpe, “Multitone Direct Sequence CDMA System in an IndoorWireless Environment”, IEEE First Symposium on Communications andVehicular Technology; Benelux Delft Netherlands, pp.4.1-1 to 4.1-8, Oct.27-28, 1993; and

K. Fazel, “Performance of CDMA/OFDM for Mobile Communication System”,2nd IEEE International Conference on Universal Personal Communications,Otawa, Ontario, pp.975-979, Oct. 12-15, 1993.

The following references describe various methods to combine adaptivebeamforming with processing the spreading codes in CDMA:

G. Tsoulos, et al. “Adaptive Antennas for third generation DS-CDMAcellular systems”, Proc. IEEE VTC'95, pp.45-49, August 1995.

Y. Wang et al., “Adaptive antenna arrays for cellular CDMA communicationsystems”, Proc. IEEE Intl. Conf. Acoustics, Speech and SignalProcessing, Detroit, pp. 1725-1728, 1995.

B. Quach, et al, “Hopfield network approach to beamforming in spreadspectrum communications”, IEEE Proc. Seventh SP Workshop on StatisticalSignal and Array Processing, pp. 409-412, June 1994.

A. Sandhu, et al. “A Hopfield neurobeamformer for spread spectrumcommunications”, Sixth IEEE Int. Symposium on Personal, Indoor andMobile Radio Communications, September 1995(no page given)

A. F. Naguib, et al. “Performance of CDMA cellular networks withbase-station antenna arrays”, in C. G. Gunther, ed. “MobileCommunications—Advanced systems and components”, Springer-Verlag, pp.87-100, March 1994.

V. Ghazi-Moghadam, et al, “Interference cancellation using adaptiveantennas”, Sixth IEEE Int. Symposium on Personal, Indoor and MobileRadio Communications, pages 936-939, September 1995.

H. Iwai, et al. “An investigation of space-path hybrid diversity schemefor base station reception in CDMA mobile radio”, IEEE J.Sel.Areas,Comm., vol.SAC-12, pp.962-969, June 1994.

R. Kohno, et al. “A spatially and temporally optimal multi-user receiverusing an array antenna for DS/CDMA”, Sixth IEEE Int. Symposium onPersonal, Indoor and Mobile Radio Communications, Toronto, pages950-954, September 1995.

Despite these suggestions to combine certain of the multiple accessprotocols to improve bandwidth efficiency, there has been little successin implementing such combinations. One reason for this lack of successis that it becomes more difficult to calculate optimum operatingparameters as more protocols are combined. The networks implementingcombined multiple access protocols become more complex and expensive.Accordingly, the implementation of high-bandwidth efficiencycommunications using a combination of multiple access protocolscontinues to be a challenge.

SUMMARY OF THE INVENTION

In accordance with this invention, a highly bandwidth-efficient methodfor transmitting and receiving information is implemented. In one aspectof the invention, a plurality of multiple-access, bandwidth-efficientcommunication techniques are combined. The invention is based, in part,on Applicants' realization that the distinct spectral and spatialanalyses of received signals may be combined in a unified operation toextract each user's signal in a highly bandwidth-efficient multipleaccess system.

One aspect of the invention is a method of communicating signals from atleast two different spatially separated remote transmitters to areceiving base station having a multi-element antenna array. Each of thetransmitters transmits signals representative of different information.In accordance with this aspect of the invention, the mathematicalrepresentation of the spectral characteristics of the signals is capableof being put in a mathematical form that is substantially the same asthe mathematical representation of the spatial characteristics ofsignals received by a multi-element antenna array. This enables thereceiver to efficiently process the received signals to simultaneouslyobtain adaptive spectral and spatial despreading and spreading weightsthat enhance the signal to noise and interference ratio of the signals.The receiver can then identify the data associated with each of thesignals transmitted by the transmitters and can forward that data to therespective recipients. The term “spreading gains” can be used instead of“spreading weights”, to emphasize the meaning that their values areadaptive and can vary in magnitude.

In another aspect of the invention, the spectral format of the signalsis what we call discrete multitone stacked carrier (DMT-SC). In thisformat, the user's data signal is modulated by a set of weighteddiscrete frequencies or tones. The weights are spreading codes thatdistribute the data signal over a plurality of discrete tones covering abroad range of frequencies. The weights are complex numbers with thereal component acting to modulate the amplitude of a tone while thecomplex component of the weight acts to modulate the phase of the sametone. Each tone in the weighted tone set bears the same data signal.Plural users at the transmitting station can use the same tone set totransmit their data, but each of the users sharing the tone set has adifferent set of spreading codes. The weighted tone set for a particularuser is transmitted to the receiving station where it is processed withdespreading codes to recover the user's data signal. For each of thespatially separated antennas at the receiver, the method of theinvention involves the transformation of the received multitone signalsfrom time domain signals to frequency domain signals. Despreadingweights are then assigned to each frequency component of the signalsreceived by each antenna element. Values of the despreading weights arethen determined which, when combined with the received signals, resultsin an optimized approximation of individual transmitted signalscharacterized by a particular multitone set and transmitting location.

In another aspect of the invention, the spectral portions of thedespreading weights are adaptively adjusted in value at the receivingstation to improve the quality of the received signal. This process isreferred to as adaptive code nulling. When the spreading codes used tospread distinct data signals are orthogonal, interference in the channelcan be removed from the spread data. However, when the spreading codesare not orthogonal, which may be the case with spreading codes that areused in neighboring spatial cells, cross modulation may result so thatthe data signals are not able to be precisely distinguished by simpledespreading. In order to compensate for this phenomenon, code-nullingweights are multiplied by the received signal. By nulling out the crossmodulation present in the received signal, the appropriate values of thedata bits are output by the receiver. The adaptive code nullingprocedure may be implemented during the derivation of the overalldespreading weights that maximize the signal quality.

In another aspect of the invention, the spatial portions of thedespreading weights are adaptively adjusted in value at the receivingstation so that the spatial directions of low gain or the null regionsof the receiver are adaptively positioned in a pattern so that the nullsare directed towards known interfering signal sources. In this manner,interfering signals are de-emphasized in the spatial domain. This “nullsteering” procedure may also be implemented during the derivation of theoverall despreading weights that maximize the signal quality.

In another aspect of the invention the mathematical formalism used toachieve null-steering is found to be analogous to the formalism used toachieve code-nulling. According to this analogy, just as the tones in atone set are multiplied by complex weights to alter the amplitude andphase of the tones, so are the gain and relative phase of signalsreceived by the antenna elements altered by a set of multiplicativeweights. This multiplication by complex weights can be expressed in amatrix form for both code nulling, which is a spectral concept, and nullsteering, which is a spatial concept. Thus, the calculations performedin the spectral code domain correspond formally to the calculationsperformed in the spatial domain. Consequently, in this aspect of theinvention null steering can be performed in a system using code-nullingsimply by adding extra “spatial” dimensions to the spectral matricesused for calculating the complex weights and multiplying the signals bythese “unified spatial/spectral” weights.

In another aspect of the invention, the signals are transmitted in timedivision duplex format—e.g. a format in which base to remote signals aretransmitted in different time periods than remote to base signals. Inone embodiment of this aspect of the invention, a first plurality ofreceived signal bursts separated by first-burst time-guard time-periods,are received by the receiving station. This is followed by transmissionby the receiving station of a second plurality of transmitted signalbursts separated by second-burst time-guard time-periods. The first andsecond bursts are separated by an interburst time-guard time-period thatis larger than either the first-burst time-guard time-period or thesecond-burst time-guard time-period. The interburst time-guardtime-period is sufficiently large to reduce interference between signalsreceived by the receiving station and signals transmitted by otherreceiving stations.

Another aspect of the invention is the application to wirelesscommunications of the mathematical analogy that may exist between therepresentation of data that has been spectrally processed in accordancewith certain multiple access techniques and data that has been spatiallyprocessed by a multi-element adaptive antenna array. Applicants findthat because of this analogy these analyses can be combined in a unifiedmathematical operation. This greatly simplifies the calculation ofoptimal operating parameters, including spreading codes, and permitsidentification of the signals associated with each individual user.Accordingly, in this aspect of the invention the dynamic, real-timecalculation of the most desirable operating parameters forhigh-bandwidth efficiency, and the identification of each user's signal,becomes more economical, despite the fact that a plurality of multipleaccess techniques are used.

In one aspect of the invention, applicants show that spreading a signalover a set of weighted tones in DMT-SC is one multiple-accessspectral-processing format that resembles the format of data that isprocessed by a multi-element adaptive antenna array. Accordingly, in anembodiment of the invention, space division multiple access (SDMA) usingmulti-element adaptive antenna array techniques is combined with DMT-SCto obtain significant calculational advantages.

In still another aspect of the invention these techniques are combinedwith higher order modulation formats, such as higher order QAM or M-aryPSK or FSK to obtain further bandwidth efficiencies.

In embodiments of the invention, a single matrix-calculation implementsthe spreading/despreading functions, including what would previouslyhave been the separate steps of code determination, code nulling, beamforming, and null steering. This operation yields despreading weightsthat result in optimum ratios of signal to noise and interference.

Currently, the invention has advantageous applications in the field ofwireless communications, such as cellular communications or personalcommunications, where bandwidth is scarce compared to the number of theusers and their needs. Such applications may be effected in mobile,fixed, or minimally mobile systems. However, the invention may beadvantageously applied to other, non-wireless, communications systems aswell.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a tutorial diagram illustrating an example of pure spectraldiversity, showing how a receiver distinguishes two sets of discretemultitone signals from two transmitters that are placed close to oneanother, in accordance with the invention.

FIG. 1B is a tutorial diagram illustrating an example of pure spatialdiversity, showing how a receiver distinguishes two discrete monotonesignals from two transmitters that are placed far from one another, inaccordance with the invention.

FIG. 1C is a tutorial diagram illustrating an example of both spectraland spatial diversity, showing how a receiver distinguishes two discretemultitone signals from two transmitters that are placed far from oneanother, in accordance with the invention.

FIG. 1D is a high-level schematic representation of an implementation ofthe invention in a fixed wireless communication system.

FIG. 2 is a simplified representation of multitone transmission.

FIG. 3 is a simplified representation of the use of a discrete multitonestacked carrier signal format.

FIG. 4 is a simplified representation of the matrix formalism used in animplementation of the invention.

FIG. 5 is a simplified representation of the matrix formalism, used inan implementation of the invention, that includes the effects of channelresponse.

FIG. 6 is a simplified representation of DMT-SC using an exemplaryhigher order QAM modulation format.

FIG. 7 is a timing diagram that illustrates the general time divisionduplex signal and protocol used in an embodiment of the invention.

FIG. 8 is a signal processing flow diagram that depicts the main signalprocessing steps used in an embodiment of the invention to provide forhigh bandwidth efficiency.

FIG. 9 is a signal processing flow diagram that illustrates a methodused to spread the encoded carrier signal.

FIG. 10 is a three-dimensional plot of the signal to interference plusnoise ratio versus code weights and spatial weights applied to thetransmitted and received signals.

FIG. 11 is a perspective cut away view showing an embodiment of a basestation antenna.

FIG. 12 is a perspective cut away view showing a second embodiment of abase station antenna.

FIG. 13 graphically depicts the null steering aspect of the presentinvention.

FIG. 14 is a schematic representation of an inverse frequencychannelized spreader implementation.

FIG. 15 is a schematic representation of a frequency channelizeddespreader implementation.

FIG. 16 is a plot of antenna gain versus angular direction.

FIG. 17 is a highly simplified block diagram that illustrates oneparticular application of the highly bandwidth-efficient communicationsnetwork of the present invention.

FIG. 18 is a list of the possible operational frequency bands of aspecific embodiment of the invention.

FIG. 19 shows the RF Band/Sub-band organization of the airlink of aspecific embodiment of the invention.

FIG. 20 shows the tones within each sub-band of a specific embodiment ofthe invention

FIG. 21 shows the traffic partitions in a specific embodiment of theinvention

FIG. 22 shows the tone mapping to the ith traffic partition

FIG. 23 shows the overhead tone Mapping to Channels for the ith Sub-bandPair

FIG. 24 shows the Division of Tone Space to Traffic and Overhead Tones

FIG. 25 shows the time Division Duplex format for Base and Remote UnitTransmissions

FIG. 26 shows Details of the Forward and Reverse Channel Time Parameters

FIG. 27 shows the TDD Parameter Values

FIG. 28 shows the Physical Layer Framing Structure

FIG. 29 shows the Phase A Sub-band Pair Assignment Within a Spatial cell

FIG. 30 shows the Phase-A Sub-band Pair Assignment Across Spatial cells

FIG. 31 is a Functional Block Diagram for the Upper Physical Layer ofBase Transmitter for High Capacity Mode

FIG. 32 is a Data Transformation Diagram for the High Capacity ForwardChannel Transmissions

FIG. 33 is a Functional Block Diagram for the Upper Physical Layer ofBase Transmitter for Medium Capacity Mode

FIG. 34 is a Data Transformation Diagram for the Medium Capacity ForwardChannel Transmissions

FIG. 35 is a Functional Block Diagram for the Upper Physical Layer ofBase Transmitter for Low Capacity Mode

FIG. 36 is a Data Transformation Diagram for the Low Capacity ForwardChannel Transmissions

FIG. 37 is a representation of the Triple DES Encryption Algorithm

FIG. 38 depicts a Feed Forward Shift Register Implementation of Rate3/4, 16PSK Trellis Encoder for High Capacity Mode

FIG. 39 depicts a Feed Forward Shift Register Implementation of Rate3/4, 16QAM Trellis Encoder for High Capacity Mode

FIG. 40 shows the Signal Mappings for Rate 3/4, 16QAM and 16PSK TrellisEncoding Schemes Employed in High Capacity Mode

FIG. 41 shows the Signal Mappings for Rate 3/4, Pragmatic 16 QAM and 16PSK Trellis Encoding Schemes Employed in High Capacity Mode

FIG. 42 depicts a Feed Forward Shift Register Implementation of Rate2/3, 8PSK Trellis Encoder for Medium Capacity Mode

FIG. 43 depicts a Feed Forward Shift Register Implementation of Rate 2/38QAM Trellis Encoder for Medium Capacity Mode

FIG. 44 shows the Signal Mappings for Rate 2/3, 8QAM and 8PSK TrellisEncoding Schemes Employed in Medium Capacity Mode

FIG. 45 shows the Signal Mappings for Rate 2/3, 8QAM and 8PSK TrellisEncoding Schemes Employed in Medium Capacity Mode

FIG. 46 depicts a Feed Forward Shift Register Implementation of Rate 1/2Convolutional Encoder for Low Capacity Mode

FIG. 47 shows the Signal Mapping for Rate 1/2, QPSK Pragmatic TrellisEncoding Scheme Employed in Low Capacity Mode

FIG. 48 shows the Gray-Coded Mapping for Rate 1/2, QPSK PragmaticTrellis Encoding Scheme Employed in Low Capacity Mode

FIG. 49 shows the Base Mapping of Elements of Received Weight Vectors toAntenna Elements and Tones

FIG. 50 is a Block Diagram Representation of CLC Physical Layer Format

FIG. 51 shows the QPSK Signal Mapping for the CLC Channel

FIG. 52 is a representation of the CLC Interleaving Rule

FIG. 53 shows the Tone Mapping of (4×4) Interleaved Matrix Elements

FIG. 54 is a Block Diagram Representation of BRC Physical Layer Format

FIG. 55 shows the Tone Mapping of the (4×4) Interleaved Matrix Elements

FIG. 56 is a representation of a Broadcast Channel Beam Sweep

FIG. 57 is a Functional Block Diagram of the Upper Physical Layer ofRemote Unit Transmitter for High Capacity Mode

FIG. 58 is a Data Transformation Diagram for the High Capacity ReverseChannel Transmissions

FIG. 59 is a Functional Block Diagram for the Upper Physical Layer ofRemote Unit Transmitter for Medium Capacity Mode

FIG. 60 is a Data Transformation Diagram for the Medium Capacity ReverseChannel Transmissions

FIG. 61 is a Functional Block Diagram for the Upper Physical Layer ofRemote Unit Transmitter for Low Capacity Mode

FIG. 62 is a Data Transformation Diagram for the Low Capacity ReverseChannel Transmissions

FIG. 63 shows the Remote Unit Tone Mapping of Received Weight VectorElements

FIG. 64 is a Block Diagram Representation of the CAC Physical LayerFormat

FIG. 65 shows the BPSK Signal Mapping for the CAC Channel

FIG. 66 depicts the CAC Interleaving Rule

FIG. 67 shows the Tone Mapping of the (8×2) Interleaved Matrix Elements

FIG. 68 is a Functional Block Diagram for the Lower Physical Layer ofBase Transmitter

FIG. 69 shows Tone Mapping into DFT Bins

FIG. 70 shows Tone Mapping into DFT Bins

FIG. 71 is a block diagram that illustrates the main structural andfunctional elements of the bandwidth on demand communications network ofthe present invention.

FIG. 72 is a functional block diagram that illustrates the mainfunctional elements of the high bandwidth remote access station.

FIG. 73 is a functional block diagram that shows the main functionalcomponents of the high bandwidth base station.

FIG. 74 is an overall system schematic block diagram that shows the mainstructural and functional elements of one implementation of the highlybandwidth-efficient communication system in greater detail.

FIG. 75A depict the digital architecture within an exemplary remoteaccess terminal.

FIG. 75B depict the digital architecture within an exemplary remoteaccess terminal.

FIG. 76 is a software block diagram that indicates the generalprocessing steps performed by each of the digital signal processingchips within the digital signal processing architecture of FIGS. 75A and75B.

FIGS. 77A-77D are block diagrams that show in detail the digitalarchitecture of the LPA cards of FIGS. 75A and 75B.

FIGS. 78A-78C are block diagrams that detail the digital architectureused to support the main digital signal processing chips on theinterface card of FIGS. 75A and 75B.

FIGS. 79A-79D are a schematic block diagram that depicts the overalldigital signal processing architectural layout within an exemplary basestation of the present invention.

FIG. 80 is a schematic block diagram showing a dual band radio frequencytransceiver that may advantageously be used in the high bandwidth remoteaccess station shown in FIG. 74.

FIG. 80A is a schematic block diagram showing the main internalfunctional elements of the synchronization circuitry shown in FIG. 80.

FIG. 81 is a schematic block diagram depicting a dual band radiofrequency transceiver that may advantageously be implemented within thehigh bandwidth base station shown in FIG. 74.

FIG. 81A is a simplified schematic block diagram showing the maininternal components of the frequency reference circuit shown in FIG. 81.

FIG. 82 is a schematic block diagram of a dual band radio frequencytransmitter of a type that may advantageously be implemented within abase station constructed in accordance with the present invention.

FIG. 83 depicts the bandwidth allocation method performed by thebandwidth demand controller of FIG. 74.

FIG. 84A and FIG. 84B show an alternate embodiment of the invention,where the spectral processing and the spatial processing are separated.

FIG. 85 is an illustrative flowchart of an embodiment of the adaptivesolution of spectral and spatial weights.

FIG. 86 is a block diagram of a plurality of remote stations coupled toa base station over a wireless link using discrete multitone spreadspectrum communication and incorporating the principles of the presentinvention.

FIG. 87 is a block diagram of a base station included in FIG. 86.

FIG. 88 is a flow diagram which implements the operation of theinvention of FIGS. 86 and 87.

FIG. 89 is an architectural diagram of the PWAN system, including remotestations transmitting to a base station.

FIG. 90 is an architectural diagram of the remote station X as a sender.

FIG. 91 is an architectural diagram of the base station Z as a receiver.

FIG. 92 is a more detailed architectural diagram of the priority messageprocessor 204 at the sending station.

FIG. 93 is a flow diagram showing the remote station as the sender andthe base station as the receiver.

FIG. 94 is a more detailed architectural diagram of the priority messageprocessor 320 at the receiving station.

FIG. 95 is an architectural diagram of the PWAN system, showing the basestation polling a remote station over the common link channel.

FIG. 96 is an architectural diagram of the PWAN system, showing theremote station transmitting a functional quality and maintenance messageto the base station over the common access channel.

FIG. 97 is an architectural diagram of the remote station X as a senderof functional quality and maintenance data.

FIG. 98 is an architectural diagram of the base station Z as a receiverof functional quality and maintenance data.

FIG. 99 is a flow diagram of the sequence of operational steps for theinvention.

FIG. 100 is an architectural diagram of the personal wireless accessnetwork (PWAN), showing the base station Z transmitting forward pilottones with a prearranged initial forward signal power level, to theremote station X and to the remote station Y.

FIG. 101 is an architectural diagram of the personal wireless accessnetwork (PWAN) of FIG. 100, showing the remote station X transmittingreverse pilot tones with a prearranged initial reverse signal powerlevel, to the base station Z.

FIG. 102 is a network diagram of two cells engaging in a first stage ofretrodirective coupling, where the base station B1 in cell 1 detects thepresence of interfering signals from the remote station R2 in theneighboring cell 2. Base station B1 adjusts its transmissions in thedirection of remote station R2 to diminish their signal strength.

FIG. 103 is a network diagram of the two cells of FIG. 102 in a secondstage of retrodirective coupling, where the base station B2 in thesecond cell 2 detects the presence of interfering signals from theremote station R1′ in the first cell 1. Base station B2 adjusts itstransmissions in the direction of remote station R1′ to diminish theirsignal strength.

FIG. 104 is a network diagram of the four cells similar to FIGS. 1A and1B, showing propagation of channel optimization across cell boundariesto optimize the channel characteristics throughout the entire system.

FIG. 105 is a more detailed block diagram of base station B1 and remotestation R1′ in cell 1 and remote station R2 in cell 2, where remotestation R2 is sending interfering signals to the base station B1.

FIG. 106 is a detailed block diagram similar to FIG. 105, showing thebase station B1 sending diminished strength signals across the cellboundary, in the direction of the interfering remote station R2.

DETAILED DESCRIPTION OF THE INVENTION

In what follows, aspects of the principles of the invention will bediscussed in a tutorial illustrating an example of pure spectraldiversity, an example of pure spatial diversity, and an example of mixedspectral and spatial diversity. This will be followed by a discussion ofthe invention in a high-level overview that will include an explanationof the waveform used in the practice of an aspect of this invention.This will be followed by a description of more specific “details of theinvention,” and then by a detailed description of a “specific embodimentof the invention.”

HIGH LEVEL OVERVIEW OF THE INVENTION Introduction

This invention is based, in part, on the realization that there is ananalogy between the mathematical description of beams formed bymulti-element adaptive, or phased, antenna arrays and the mathematicaldescription of signals that are formatted according to certain multipleaccess schemes, such as the exemplary DMT-SC. Based on this realization,applicants have been able to simplify the calculations necessary when aplurality of multiple access techniques are combined. Using thisinvention, one may more effectively use a limited bandwidth region ofthe electromagnetic spectrum to service a large number of users.Techniques that may be combined in accordance with the teachings of thisinvention include SDMA using multi-element antenna arrays, DMT-SC, andhigher order modulation formats such as higher order QAM.

Tutorial Presentation of the Invention

FIG. 1A-1C are tutorial illustrations of the technology involved inaspects of this invention. FIG. 1A is a tutorial diagram illustratinghow the same spectral frequencies can be used by two differentsubscribers, Alice and Bob. Although Alice and Bob are located in thesame place and their spectral frequencies are the same, thosefrequencies are coded differently—Alice's signal is encoded with code 1and Bob's signal is encoded with code 2. Consequently, even though thetwo signals “mix”, they can be separated in box 1A.1, based on thedifferent codes, to produce separate versions of Alice's and Bob'ssignals.

FIG. 1B is a tutorial diagram illustrating how the same spectralfrequencies can be used by two different subscribers, Chuck and Dave,who, although they are using the same frequencies, are located indifferent places. Although their spectral frequencies are the same, andtheir signals “mix”, since their signals originate in two differentlocations they can be separated in box 1B to produce separate versionsof Chuck's and Dave's signals.

If Alice, Bob, Chuck and Dave all transmit at the same time using thesame frequencies, as shown in FIG. 1C, their signals might be divided,as shown in that FIGURE.

In accordance with one aspect of this invention, the signals areseparated in one step as shown in FIG. 1C. As shown in that FIGURE, thesignals are separated in box 1C.1 in a single operation. Applicant'shave shown that the mathematical similarity between the descriptions ofthe spectral and spatial aspects of the signal permit this unifiedseparation of the signals.

The Airlink

Discrete Multitone Transmission

An exemplary communication system in which the invention may beimplemented is shown in FIG. 1D. In this FIGURE, the various elementsmarked 11 are exemplary fixed remote terminals serving users, while theboxes marked 12 are the base stations associated with certain of thoseremote terminals. It should be noted that in this context the term“fixed” remote terminals applies not only to remote terminals that donot move during use, but may also apply to terminals that are mobile, solong as they are serviced by one base station during a call. Other“fixed” embodiments may permit motion between spatial cells during acall and motion at less than 10 or 5 miles per hour. Additional remoteterminals and base stations are also shown.

The remotes and base stations are connected by exemplary airlinks, 13.The base stations may be connected to a “wireless network controller”14, which then connects to the wider telecommunications network, 15.Connections between the base station and the network controller andbetween the controller and the telephone network, may be wired orwireless.

An aspect of the invention is centered about the exemplary airlinks, 13,that connect the base stations and the remotes. These airlinks usescarce bandwidth resources and are advantageously operated in a highlybandwidth-efficient mode so as to accommodate a large number of users.The airlink, shown as 13 in FIG. 1D, involves a multitude of complextransmission techniques. A first exemplary technique is an embodiment ofmultitone transmission that we call “Discrete Multitone Stacked Carrier(DMT-SC)”. In this technique, a signal is transmitted over discretecarrier frequencies, shown in FIG. 2 as 21. Specific tones can beassigned to specific users, as shown in the FIGURE. As discussed above,the tones may be spaced at a frequency of 1/T, where T is the symbolrate, so that they are “orthogonal”—i.e., they do not interfere with oneanother—as in OFDM. Each tone can carry different data, but for thepurposes of this discussion, at least some of the various tones assignedto a specific user will be assumed to be carrying redundant informationto realize the advantages of frequency diversity. Such redundanttransmission over a range of frequencies allows recovery of the signaleven if some frequencies are subject to severe interference—a problem ofparticular interest in the embodiment of this invention that involvesfixed remotes. As mentioned above, certain implementations of thissignal format enables analysis that can be effected using fast Fouriertransform calculational techniques

DMT-SC

In an aspect of this invention, bandwidth efficiency is increased byspreading the signal over a set of weighted tones, with each user beingassigned a specific set of tones and weights. This technique, isdepicted in FIG. 3. In this FIGURE, identical data is sent over the fourtones identified as 1, 2, 3, and 4. User 1 is to be sent a “+1.” User 2is to be sent a “+1.” User 3 is to be sent a “+1.” User 4 is to be senta “−1.” The same tones are used to send information to the fourdifferent users by using different “weights” for each user. Theseweights may be viewed as user-specific codes, and we may refer to themas weights, codes or weight-codes. In this heuristic example, theamplitude of a particular tone is obtained by multiplying the data valueby the weight-code value for that combination of user and tone. Forexample, the weight-code of the second user is [1 −1 1 −1], meaning thatfor the second user the amplitude of the first tone is the data valuetimes +1, the value for the second tone is the data value times −1, etc.For example, the value of the second tone for the second user is thedata value, +1, multiplied by the weight-code value for the seconduser's second tone, −1, to yield a −1, as shown in the second positionof the second line. This process is called “spreading” since it effectsthe spreading of the data across the tone set.

The various tone values are added to obtain the composite spectrum,shown on the last line of the FIGURE, that is then transmitted. Uponreceipt of the spread data, the data is “despread”, i.e., the data to besent to the various users is obtained by multiplying the compositespectrum by the inverse of a particular user's weight-code. This can beperformed simultaneously for all users by using appropriate matrixtechniques.

It is helpful to bear in mind the difference between this “discretemultitone stacked carrier (DMT-SC)” technique and well known embodimentsof the classical spread spectrum technique. In DIRECT SEQUENCE SPREADSPECTRUM, each data symbol is multiplied by a series of code pulses.This spreads the data over a much wider region of the spectrum. InFREQUENCY HOP SPREAD SPECTRUM, the data is transmitted over differentregions of the spectrum during different time slots, in accordance witha pre-defined hopping code. In the DMT-SC used in this invention, thesignal is modulated by a set of weighted discrete frequencies, not overa continuous broad frequency range, as in direct sequence.

It should be appreciated, that although depicted in this example as aset of real numbers, the spreading codes advantageously comprise avector wherein each vector is a complex number.

Matrix Representation of the “Coding” Process

Exemplary, high level equipment arrangements used in the “spreading” and“despreading” is represented schematically in FIG. 4. In this FIGURE, 41is the data, D, that is to modulate a DMT-SC signal. At 42, the variousDMT-SC carriers are encoded as depicted previously in FIG. 3. Themathematical description of the spreading operation is shown in formula43, where SD is the “spread data,” CM is the “code matrix,” and D is the“data.” The detailed matrix operation is shown in formula 44, where 45is the data vector array representing the data of FIG. 3, 46 is the codematrix of that FIGURE, and 47 is the composite spectrum or spread datavector array.

When the spread data, SD, is received, it is “despread” by means of thevector operation shown in 60, where SD is the received “spread data,”CM⁻¹ is the inverse of the code matrix and DD is the “despread data,”that, as required, is reflective of the original data. This vectoroperation is shown in detail in 48, where 49 is the received spreaddata, 50 is the inverse of the code matrix, and 51 is the despread data.It is important to note for the discussion in the next section on theeffects of SDMA, that the size of this code matrix is determined by thetotal number of tones that are used.

As will be discussed below in the section on “Specific Details of theInvention,” it is not necessary to use orthogonal codes. In fact, inmost embodiments of this invention the codes are usually only linearlyindependent, and the effects of cross talk between users with differentcodes is treated using a “code nulling” process that resultsautomatically from the practice of one aspect of this invention.

Use of SDMA in “Coded” Multitone Transmission

An important aspect of the invention involves the realization that themathematical description of a certain processed spectrally processedsignals, such as DMT-SC signals, is analogous to the mathematicaldescription of a signal spatially processed by a multi-element adaptiveantenna array. Accordingly, the mathematical description of suchspectrally processed signals may simultaneously describe spatialprocessing by a multi-element adaptive antenna array by simplyincreasing the size of the DMT-SC matrix to take into account the numberof antenna elements in the antenna array. The dimensionality of thecombined “spectral/spatial matrix” that comprises the spreading weightsby which each tone is multiplied, is then equal to the number of tonesmultiplied by the number of energized antenna elements.

As noted in the Summary of the Invention, the mathematical formalismthat describes an aspect of this invention treats both code and antennaaspects of the received signal similarly. The signal processing maytherefore automatically result not only in codes that have minimal crosstalk with other coded signals, but also in the formation of beams thatyield minimal interference from users illuminated by different beams.These advantages are usually derived separately and are known as codenulling and null steering respectively. They will be explained ingreater detail below in the section on “Specific Details of theInvention.”

Channel Response, Equalization, and Signal Extraction

The discussion to this point has not involved any description of theeffects of channel response. Distortions due to the channel response canbe introduced into the formalism by means of a “channel response”matrix, shown in FIG. 5. In this FIGURE, the received data “RD”, shownin 52, is no longer equal to the spread data as in FIG. 4, but is nowdistorted by a channel response “CR.”. The received data is then theproduct of the spread data and the channel response, as shown in 52.This effect is shown in 53, with the exemplary numbers used in FIG. 3.As shown, the despread data, 54, now do not have the original datavalues, but rather values that are distorted by the channel response. Inorder to correct this distortion, the code despread matrix must includeterms that “equalize” the channel distortion.

In an embodiment of the invention, this channel response vector isdetermined by transmitting a pilot signal and noting its distortion bythe channel (“pilot driven equalization”). In another embodiment, theeffect of channel response is “equalized” by simply adaptivelycalculating a “despread matrix” that maximizes the ratio ofsignal-to-noise-and-interference associated with the transmitted data(data driven equalization). The calculated optimum system parameters mayinclude a mathematical representation of the channel response. Thesechannel response parameters may then be used by either the base or theremote to “equalize” the channel distortion. These parameters may beused by the either side of the link because, at least for short periodsof time, the channel is reciprocal in time. Of course, thesecalculations may be equally well done at some more central locationrather than at the remote. What is central to this aspect of theinvention is that certain calculations can be reused. Accordingly,despread weights used on the receive signals can be reused, with onlyminimal modifications, to spread signals on the next transmission—aprocess called retrodirectivity. Additionally, it may be possible toreuse, at the remotes, the weights that are calculated at the basestation.

Of course, the calculation of optimum system parameters, and theadaptive extraction of the data associated with each user from thecombined signal, is done taking into account all of the system signalsthat are seen by the base and transmitted by the base. Accordingly, theremotes may calculate their own despread weights to take into accountthe interfering signals that they receive, which are not received by thebase station. Simplification may be achieved by using a fixed beampattern for the remotes, rather than calculating beam weights, since theremote knows where the fixed base is and need not reoptimize its beamweights once its beam pattern is fixed.

Modulation Formats

Although, to this point, we have shown the signal as either zeros orones, it is clear that the carrier may be modulated in any one of anumber of signal modulation formats, such as higher order QAM. In suchan exemplary format, the composite spectrum will be as shownheuristically in FIG. 6. In this FIGURE, the tone sets, as shown in FIG.2, are displayed on the x-axis. The different codes used for the varioustone sets are shown on the z-axis. Finally, the constellationsassociated with QAM modulation are shown on the y-axis as circles. Theparticular constellation point that is energized is represented by aclosed circle. The composite spectrum, obtained by “collapsing” thez-axis, is shown x-z dimension. The blur represents the composite of allof the energized constellations of a particular tone.

Time Division Duplex

In an embodiment of the invention, the bandwidth-efficient transmissiontechniques used in the invention are combined in a Time Division Duplex(TDD) configuration, i.e., a configuration in which the channel isdivided into time slots with uplink and downlink transmission occurringalternately in adjoining time slots. A simplistic TDD configuration isshown in FIG. 7. As shown in the FIGURE, during alternate time slots,information is sent uplink (from base station to remote) and thendownlink (remote to base station). The guard time is selected to allowfor the delay time due to multi-path. All remotes and base stations maybe synchronized so that all remotes transmit at the same time and thenall base stations transmit at the same time. Well known GPS techniquesmay be used for such synchronization.

As indicated above, the use of TDD, and the assumption of a channelresponse that varies slowly compared to a TDD period, permits theinterchangeable use of spread and despread weights, at least duringcontiguous receive and transmit cycles at a given location. Likewise itmay be possible to reuse, at a second location, the larger part of the“spread/despread matrix” calculated at a first location. For example, itmay be possible to use at the remotes the weights—from which each usersinformation can be extracted—that were calculated at a base stationduring a previous TDD period by maximizing thesignal-to-noise-and-interference ratio for the signals received at thebase. In this embodiment, the base sends to its remotes theirappropriate “channel equalized codes” or “weights” using the TDD format.The remotes may perform at least some weight recalculation, but may relyon some of the weight analysis performed at the base station. In thisway the larger part of the calculation may be done at the base stationsor at some other location removed from the remote locations. Thisreduces the cost and complexity of the more numerous remotesconsiderably.

Of course, to implement this alternate embodiment of the invention,optimization parameters must be relatively constant during the timeperiod of one uplink and one downlink time slot. Thereafter newlycalculated optimization parameters may be determined, and sent to theremotes, on a periodic basis.

Bandwidth on Demand

As noted above, the invention is particularly well adapted to providingvariable bandwidth on demand. The provision of such additional bandwidthis effected by simply assigning more tones or codes to the requestinguser, or by transmitting in a higher order modulation format.

EXEMPLARY DETAILS OF THE INVENTION

The Analogy Between DMT-SC and Adaptive Antenna Array Processing

The aspect of this invention that involves the use of spectral multipleaccess techniques that are mathematically analogous to the mathematicaldescription of adaptive antenna array signals can be better understoodin the context of heuristic FIG. 14. In that FIGURE, 10 is themathematical description of Discrete Multitone Stacked Carrier (DMT-SC).In 10, a baseband signal, d(t) is multiplied by a spreading codecomprising a set of tone frequencies and associated carrier weights,g_(k). It should be appreciated that this is different than DirectSequence Spread Spectrum where the baseband signal is multiplied by a PNcode, rather than by a set of weighted carriers. The expression of 10can be rewritten in block form, as in 20. Here the weighting operationassociated with g_(k) has been separated from the exponential operation,that we characterize in the FIGURE as an “inverse frequencychannelizer”, for example an inverse FFT. In a point central to thisaspect of the invention, applicants have recognized that thisrepresentation is analogous to that of an adaptive antenna array—where abaseband signal is multiplied by an aperture vector.

The DMT-SC despreading operation is shown in heuristic FIG. 15 and italso is found by applicants to be analogous to similar expressions foradaptive antenna array processing. In 10, a wideband signal x(t) ispassed through a bandpass filter, BPF, and then multiplied by adespreading code, w(t) nominally the inverse of the spreading code ofFIG. 15, to get the original signal, d(t). By calculating the despreaderweights appropriately, for example to maximize signal to noise, we canautomatically correct for channel distortion and other interferingsignals. In 20, the despreading operation is again separated from theexponential coefficients, as in FIG. 14. Here the received signal, x(t)is represented as equal to the sum of an interference term, i(t) and thetransmitted signal, s(t) multiplied by a distortion term h(t). From FIG.14, the transmitted signal s(t) is equal to g times d(t), givingequation 30 in FIG. 15. Applicants have recognized that this equation isanalogous to that describing the output of an adaptive antenna array.

In accordance with an aspect of this invention, this analogy betweenDMT-SC and adaptive antenna array processing leads to the possibility ofcombining both spatial and spectral expressions in one mathematicalexpression that can be solved in one unified spectral/spatialcalculation. This also leads to the further discovered analogy betweennull steering in adaptive antenna arrays and code nulling in CDMA ingeneral, and in DMT-SC in particular. In accordance with another aspectof the invention, instead of setting the despread weights to previouslyestimated spectral spreading and beam steering weights, we adaptivelycalculate the despread weights to maximize some general measure ofsignal quality—either characteristics measured directly from the channelor obtained from a blind adaptive operation or a combination of the two.

Time Division Duplex

The TDD signaling protocol used in an embodiment of the airlink isdepicted in FIG. 7. It should be noted that two 5 MHz frequency bandsseparated by 80 MHz are depicted in FIG. 7. In one embodiment, the samedata is transmitted over both 5 MHz bands to reduce multi-path fadingeffects. The 80 MHz separation between the two bands ensures that thesame multi-path fade does not interfere with both bands. In addition,the 5 MHz frequency band is divided into four 1 MHz sub-bands, with thelower and upper 500 Hz of each 5 MHz band being designated as guardbands. The four 1 MHz sub-bands in the lower 5 MHz band are matched withcorresponding sub-bands in the upper 5 MHz band. So, for example, thefirst sub-band in the lower 5 MHz band shares is duplicated in the firstsub-band in the upper 5 MHz band, etc.

In one preferred embodiment, the transmission period from the basestation to the remotes and from the remotes to the base station(T_(symbol)) is approximately equal to 340 microseconds. The guard time(T_(guard)) between transmission and reception is approximately 35microseconds, in one embodiment, while the total revisit time(T_(revisit)) is approximately 750 microseconds. As has been mentioned,and as will be discussed in greater detail below, it is important thatT_(revisit) be less than the amount of time in which significant changesare likely to occur over the air channel, in order to ensure thatessentially the same channel characteristics are observed within anyselected interval of length T_(revisit).

The guard time, T_(guard), between the forward and reverse bursts mustbe sufficiently large to allow significant attenuation of multi-pathreflections. Base-to-base interference necessitates sufficient guardtime between forward and reverse transmission bursts. In an embodimentof the invention four forward bursts are transmitted without anyintervening reverse bursts and then four reverse bursts are transmitted.The guard time between the four bursts is much smaller than the guardtime at the end of the four bursts. This reduces the base-to-baseinterference.

High Level View of the Signal Processing

FIG. 8 is a signal flow diagram that generally illustrates the signalprocessing steps performed in one embodiment of the invention on anaudio, video, voice or data signal transmitted over the air interfacebetween the base station and the remotes. As shown in FIG. 8, a signal(that may comprise audio, video, voice or data) is supplied from acommunication link to an input terminal 1010. This signal is thenpacketized in digital format as indicated by a block 1011. The signal ispacketized so that the entire signal can be sent in a single packetduring the transmission time T_(packet). As indicated within block 1012,the packetized signal is subsequently quadrature amplitude modulation(QAM) encoded and error encoded (using, for example, well knownReed-Solomon and/or trellis encoding techniques). Of course, it shouldbe understood that in other advantageous embodiments of the invention,binary phase shift keying (BPSK) or M-ary phase shift keying (MPSK) maybe employed as an alternative modulation technique to QAM.

The mapper 1012 outputs a complex number representative of an n-bitbinary value based upon the mapping scheme. For example, if 16 QAM isused, then the encoder 1012 will output a four-bit binary value, one of16 possible values since 2⁴=16. Likewise if 256 QAM is used, then theencoder 1012 will output one of 256 complex values representing aneight-bit binary value since 2⁸=256. The bits that enter the mapper mayhave been forward error correction encoded to protect against channelerrors.

The encoded signal is then spread over a portion of the frequency bandas indicated within a block 1013. In accordance with an embodiment ofthe invention, the DMT-SC spreading technique is used to spread theencoded signal over several frequency tones within the total frequencyspectrum. The method used to spread the encoded carrier signal isdescribed in greater detail immediately below with reference to thesignal processing flow diagram of FIG. 9.

Parallel Data Transmission Using Multitones

The division and transmission of a signal over a number ofcarriers—Parallel Transmission—is discussed, for example, in a paperentitled “Analysis and Simulation of a Digital Mobile Channel UsingOrthogonal Frequency Division Multiplexing,” by Leonard J. Cimini, Jr.,IEEE Transactions on Communications Vol. Com 33, No. 7, July 1985.Briefly, Parallel Transmission is a signal processing technique thatconverts a serial data stream into a parallel data stream, and modulatesdifferent discrete carrier tones with each of the parallel data streams.

For example, consider a set of carriers (called a tone set) thatincludes four tones. A serial data stream is then divided into fourparallel data streams by taking every fourth symbol and assigning it toa particular one of the tones. So, for example, the first, fifth, andninth symbols are assigned to the first tone; the second, sixth, andtenth symbols are assigned to the second tone, etc. Accordingly, thefirst tone in the tone set will be set to an amplitude and phasecorresponding to the symbol values output onto the first parallel datastream, the second tone in the tone set will be set to an amplitude andphase corresponding to the symbol values output onto the second paralleldata stream, etc. In a particularly advantageous embodiment of theinvention, the spacing between the tones is carefully selected toprovide orthogonal frequency division multiplexing (OFDM).

A PN code method of implementing such a modulation scheme, is depictedin FIG. 9 As indicated above, the data in the parallel data streams maybe the same or different data. The major advantage of this technique isthat it can be shown that such a processed signal is effectively theFourier transform of the original data stream, and that a bank ofcoherent demodulators is effectively an inverse Fourier transform. In anaspect of the invention, these technique is used to obtain thecalculational advantages of FFT and IFFT processing

DMT-SC Details

In an exemplary embodiment of the invention, the total bandwidthallocation for the airlink is 10 MHz. in the range of 1850 to 1990 MHz.The total bandwidth is divided into two 5 MHz bands called the Lower RFband and the Upper RF band. The separation between the lowest frequencyin the Lower RF Band and the lowest frequency in the Upper RF Band (DF)is 80 MHz. The base frequency (f_(base)) for the network is defined asthe lowest frequency of the Lower RF Band.

The Lower and Upper RF Bands are further subdivided into sub-bands. Thefirst and last 0.5 MHz of each band are designated as guard bands andare hence unused. The remaining 4 MHz in each band is then subdividedinto 4 Sub-bands sequentially numbered from 0 to 3. Each Sub-bandcontains a set of frequencies in the Lower RF Band and another set offrequencies in the Upper RF Band. The extension L indicates the setwithin the Lower RF Band and U indicates the set within the Upper RFBand.

In one embodiment there are a total of 2560 frequency tones equallyspaced in the 8 MHz of available bandwidth. There are 1280 tones in eachBand, and 640 tones in each Sub-band (320 frequencies in the lower bandand 320 frequencies in the upper band). The spacing between the tones(Df) is simply 8 MHz divided by 2560 that translates to 3.125 KHz. Thetones may be further organized into Tone Sets each with four tones, andTone Partitions, each with 20 Tone Sets. Alternatively the tones may beorganized into Tone Clusters each with 20 tones, and Traffic Partitions,each with 4 Tone Clusters. A traffic channel requires at least onetraffic partition. Control and access channels may be interspersed amongthe traffic channels in the 5 MHz slots. As will be discussed furtherbelow, data is redundant over a tone set.

The organization of the tones also permits standardization of toneassignments to users so as to permit the contemplated calculations in anorderly fashion. For example, each user may be assigned only multiplesof traffic partitions. The division of the total transmission band intosub-bands also allows for lower sampling rates and less intensive DSPrequirements (since the processed band is spread over a significantlysmaller bandwidth). In addition, the partitions provide a convenientdivision for reducing the dimensionality of received vectors. This couldbe accomplished by combining selected tone set values (i.e., thecorresponding tone set values in each cluster set). Although thisinvolves a reduction in the number of degrees of freedom, such atradeoff can be advantageous in systems wherein the maximum number ofdegrees of freedom are not necessary to accurately decode the data.Thus, by reducing the dimensionality of the tone set vector, theprocessing cost is significantly reduced.

As indicated above, to ensure that the signals modulated onto theseparate tones do not mutually interfere by overlapping with other tonesthe tones set are spaced at intervals of 1/T, the symbol rate. Ofcourse, some distortion occurs during transmission so that someinterference may occur which may be removed with additional errorcorrecting techniques.

The Use of DMT-SC

As noted above, the signals may be initially spread over assigned tonesusing appropriate codes or weights. These codes may be orthogonal withina given spatial cell, and may be randomly assigned to the same tone binswithin adjacent spatial cells. Thus, spreading codes may be reused inadjacent spatial cells and also may have a random correlation betweenadjacent spatial cells. Although the initial code assignments made bythe base station may be orthogonal, it will be understood that inresponse to the weight adjustments made during adaptive equalization,the spreading codes will typically evolve, or adapt, to non-orthogonalcodes after the communications network has been active for some time. Aswill be discussed in greater detail below, the criterion for thespreading codes used within a given spatial cell is advantageouslylinear independence rather than orthogonality. The random correlation ofspreading codes in adjacent spatial cells is compensated for by means ofan automatically implemented code nulling technique that nulls outcorrelated portions of the transmitted signals using linear weighting.

Once the spreading codes associated with the DMT-SC modulation techniquehave been assigned to the encoded data signals, as represented by theblock 1013 of FIG. 8, the processed signals are linearly summed, asindicated by a summing block 1025. A similar signal processing procedureis used on other incoming signals as indicated by the blocks 1021-1023,that correspond to the blocks 1011-1013. These signals are summed withinthe summer 1025 and assigned to carrier frequencies within the 5 MHzsub-bands shown in FIG. 7, as indicated by a block 1030.

As noted above, during the course of adaptive equalization, the codestypically become non-orthogonal in order to maximize the SINR throughoutthe overall communications network 100. However, in order to retain themaximum number of degrees of freedom throughout the communicationssystem 100, it is preferable to maintain linear independence of thecomplex spreading vectors throughout a given spatial cell. Linearlyindependent complex vectors are those that cannot be expressed as a sumor scalar multiple of any combination of the other complex vectors inthe system. Thus, by preserving linear independence among the spreadingcodes, a matrix set of linear equations can be derived that allows eachof the system variables (i.e., data symbols) to be uniquely decoded.Insofar as the spreading codes become more linearly dependent, theability to discriminate amongst data symbols becomes more difficult.However, in some applications, band pass filter values are establishedin the beginning and thereafter the system must operate within thoseconstraints.

The spread signals are linearly added on a carrier-by-carrier basis toobtain the overall DMT-SC waveform. In order to despread this signal,the received signal is detected and converted into matrix form. Thereceived vector is multiplied by a scaling factor (that is proportionalto the number of bits in the spreading code), and a matrix comprised ofthe spreading codes. The resultant vector provides the despread datasymbols as an output. From this example, it can be induced that as manydata bits can be distinctly despread as there are bits in the spreadingcodes, so long as the spreading codes remain linearly independent.

Returning to the spreading of the data, once the encoded,spread-spectrum signals have been assigned to the frequency carrierbands, the signal may be transformed from the discrete frequency domainto the analog time domain using an inverse fast Fourier transform (IFFT)and an analog to digital converter. By using an IFFT and an FFT toprovide for OFDM, multiple modulators are not required, as is well knownin the art. This is because the calculations relating to the DMT-SCmodulation technique are less intensive in the frequency domain than inthe time domain. For this reason, the bulk of the signal processing ispreferably performed in the frequency domain (with the exception of themodem operations of, for example, encryption, filtering, etc.) and istransformed to the time domain as one of the last steps beforetransmission.

Depending upon bandwidth considerations, signals from the same user areassigned one or more spreading codes and one or more traffic partitions.The assignment of different spreading codes and additional trafficpartitions to provide additional bandwidth for a requesting user unit (aunit that communicates via one of the remotes) is particularly elegant(in comparison with bandwidth allocation using TDMA) from animplementation standpoint. This is because the allocation of newspreading codes and tone sets is mathematically simple and merelyrequires a numerical change to the despreading vector (for thereassignment of a new tone set) or an increase or decrease of thebandwidth of a bandpass filter on the receiving side (for thereassignment of a new tone set).

Advantages Associated with DMT-SC

The use of DMT-SC is highly advantageous in the system of the presentinvention. For example, the use of DMT-SC allows the channelcharacteristics to be evaluated at discrete points that can be exactlyrepresented in matrix form as a complex vector. Thus, because selectedtones within each tone set can be designated as pilots distributedthroughout the frequency band, a simple evaluation of a finite number ofcomplex values results in an accurate channel estimation. Furthermore,theoretically, the channel distortion can be compensated at the discretetone frequencies by a simple complex conjugate multiplication. That is,since discrete tones are used, it is not necessary to know the entirechannel response between the tones since the channel only affectsoperations at the exact points of the tone set frequencies. If thechannel is defined at these discrete points, the received tones needonly be multiplied by the appropriate complex, amplitude and phase toequalize the channel. This means that exact equalization is accomplishedby a simple complex multiplication. This channel equalizationcalculation may be subsumed in the calculation of despread/spreadweights that improve or optimize characteristics of the signal such asthe signal to noise and interference ratio.

Also, the use of DMT-SC ensures that the equalization of antenna arraytime dispersion is very simple. In multiple element antenna arrays, atime delay is observed between receptions of a waveform by the spatiallyseparated sensors when the wave impinges on the array. In a very wideband system, this delay creates dispersion. However, by using DMT-SC,the dispersion can be represented by discrete values of a scaleablevector since the response is only evaluated at discrete points of thefrequency.

Furthermore, each user on the system could operate with a different QAM(or other M-ary) constellation size. This is because the symbols are notspread over the entire bandwidth as in direct sequence spread spectrum.Rather, in DMT-SC the symbols are spread over frequency bins of varioussizes so that each user can have the optimum size QAM constellation(i.e., the highest order allowable in a given SINR). This increases theoverall system capacity since the system is not restricted to the lowestcommon denominator (i.e., the QAM or M-ary constellation size at whichall channels can operate). In addition, at lower constellation sizes alower signal-to-noise ratio is required to demodulate the signal, andthis lower signal-to-noise ratio requirement can be used to extend therange of the base station that provides additional system flexibility.

The use of DMT-SC modulation also provides several unexpected advantageswhen used in combination with certain communication technologies. Firstof all, since DMT-SC spreading allows for flexible spreading bandwidthsand gain factors (i.e., a given signal can be spread over as muchbandwidth as desired), it is particularly advantageous for exploitingthe spectral diversity of the channel. That is, since the channel hascertain bands with better response than other bands, signals can beselectively spread over the more desirable bands.

In addition, DMT-SC also allows for the use of code-nulling to greatlyimprove the reuse capacity of the communication link beyond the reusecapacity of conventional CDMA. Since DMT-SC is used instead of directsequence or frequency hopping, selected portions of the spreading codecan be nulled within the despreader. Thus, only those portions of thespreading code which are not common with the interfering spreading codeswill be despread. Furthermore, DMT-SC is particularly advantageous whenimplemented within a variable bandwidth system since the allocation ofbandwidth is highly flexible in such a system, and can be implemented bythe appropriate assignment of additional tones to the requesting user.In summary, DMT-SC provides a solution that nulls the interferingsignals.

Finally, DMT-SC is advantageous as applied to a multi-element antennaarray system where matrix calculations comprise the bulk of theprocessing operations. As is well known in the art, as thedimensionality of a matrix grows, the calculation operations necessaryto invert the covariant matrix increases as the cube of the matrixdimensionality. Thus, the processing power increases as the cube of thematrix dimensionality and, consequently, so does the cost of theprocessing circuitry. Thus, in order to avoid skyrocketing costs, it isadvantageous to limit the dimensionality of the matrices used to performthe spreading and despreading calculations. Since in a multi-elementantenna array system it is sometimes desirable to change the number ofantenna sensor elements to enhance the beam forming capability of thesystem, such a system would normally incur an increase in matrixdimensionality (since each sensor corresponds to an element in thematrix). However, in a DMT-SC system, if sensors are added to theantenna array, the dimensionality of the matrix can be preserved byreducing the number of tones in each tone set.

This preservation of matrix dimensionality is possible because themathematical formalism used when performing amplitude and phaseweighting of the signal on each of the sensors is substantially similarto the formalism used when performing amplitude and phase weighting ofeach of the tones in a tone set. Thus, an analogy exists between themultiple sensors in an antenna array and the multiple tones in a toneset. Consequently, the same matrix can be used to determine weights forboth sensor elements and tones, so that if the number of sensor elementsincreases, the number of tones can be decreased to compensate (i.e.,preserve the same matrix dimensionality), and vice versa. Furthermore,essentially the same SINR is preserved in such a system since thedegrees of freedom lost in the number of tones is regained in the numberof beams. In contrast, direct sequence spread spectrum could not changethe number of tones as beams are added since there are no tones to addor subtract. Thus, the cost of such a system would increase enormouslyrelative to the cost of the system of the present invention as capacityis increased. Specifically, the cost of the present invention increasesapproximately proportionally with the capacity, while the cost ofanother system using, for example, direct sequence spread spectrum,increases as the cube of the capacity.

Once the signal has been DMT-SC modulated, the signal is output to theantenna for transmission. DMT-SC enables the appropriate signals to bedirected to the appropriate user units (i.e., by means of antennabeam-forming discussed below).

Beam Forming

In accordance with one aspect of the present invention, adaptive antennaarrays are used in conjunction with a beam forming algorithm to achievespatial diversity within each spatial cell and implement SDMA. That is,signals output by the antennas are directionally formed by selectivelyenergizing different antenna sensors with different signal gains so thatremote terminals in one portion of a spatial cell are able tocommunicate with the base station while other remote terminals in adifferent portion of the spatial cell may communicate with the same basestation, even if they are using the same tone set and code. It should beunderstood that in the fixed implementation of the current invention,i.e., where the remote access terminals do not move substantially duringcommunication with the base station, usually staying within a spatialcell during communication, the beam forming algorithm used in theairlink need not account for mobile remote units leaving and enteringthe spatial cell. In one advantageous embodiment, each spatial cell ispartitioned into four sectors where each sector transmits and receivesover one of the four sub-band pairs.

As set forth above, the beam forming method of the present invention,like the use of codes, should not be conceived as separate from theoverall adaptive equalization method of the present invention. Rather,the method used to selectively energize the antenna sensors (duringtransmission) or selectively weight the signals received on thedifferent sensor elements (during reception) is subsumed into theoverall method used to maximize SINR. The relation of the beam formingmethod to the overall maximization of SINR method will be described ingreater detail below.

Code-Nulling

The use of spread-spectrum technology (particularly DMT-SC) anddirectional antennas within the preferred airlink of the presentinvention allows for several error cancellation benefits, includingeffects that are analogous to code nulling and null steering, by meansof linear weighting in code and space.

Code-nulling is used to discriminate between non-orthogonal signalsemanating from adjacent spatial cells. Again, the code-nulling methodshould be understood in the context of the maximization of SINR methodof the present invention. That is, the code-nulling method should beconsidered as the portion of the method that maximizes SINR with respectto the code domain. This way of understanding the code-nulling methodwill be described in further detail with respect to FIG. 10.

It should be understood that if signals generated within the samespatial cell or beam all have orthogonal spreading codes, code-nullingis typically not necessary since the orthogonality is sufficient toensure that there is no cross modulation. However, as mentioned above,the spreading codes used within a particular spatial cell may not beorthogonal, although they are preferably linearly independent.Furthermore, the transceivers within the neighboring spatial cells mayemploy spreading codes that have a random correlation with the spreadingcodes used in the local spatial cell.

By adjusting the spreading weights associated with each communicationschannel the base station is able to cross-correlate these signals on thesame tone set to subtract out interference due to “neighboring” signals.In one embodiment, the base station has the spreading codes used tospread different signals assigned to the same tone set, so that thisinformation can be used to initially calculate the appropriate weightsfor nulling out interference from other codes.

As discussed above, when the spreading codes used to spread distinctdata signals are orthogonal, the spread data can be precisely recoveredduring despreading. However, when the spreading codes are not orthogonal(as is the case with spreading codes that are used in neighboringspatial cells), cross modulation may result so that the data signals arenot able to be precisely distinguished by simple despreading (i.e.,despreading without code-nulling).

In order to compensate for this phenomenon, code-nulling weights areused in the despreader. By nulling out the cross modulation present inthe received signal, the appropriate values of the data bits are outputby the receiver. As long as the complex spreading weights are linearlyindependent, and the SNR is sufficiently high, the exact symbol valuescan be discriminated by this method. It will be appreciated that thecode-nulling procedure above is inherently implemented during derivationof the overall weights that maximize the SINR.

Null-Steering

In addition to code-nulling, an exemplary directional antenna shown inFIGS. 11 and 12 with no spectral spreading, forms signals including nullregions (i.e., regions where the antenna attenuates incoming signals orwhere there is a very low antenna gain). These null regions can beformed in a pattern so that the nulls are directed towards knowninterferers (e.g., interfering signal sources or interfering multi-pathreflectors). In this manner, interfering signals are de-emphasized inthe spatial domain. As will be discussed in greater detail below, theuse of null-steering in conjunction with code-nulling is highlyadvantageous.

In accordance with one aspect of the present invention, significantprocessing time and sophistication can be saved since significantsimilarity exists between the methods for performing null-steering andcode-nulling. Specifically, the mathematical formalism used to achievenull-steering is analogous to the formalism used to achievecode-nulling. According to this analogy, just as the tones in a tone setare multiplied by complex weights to alter the amplitude and phase ofthe tones, so are the gain and relative phase of signals output andreceived by the antenna elements altered by a set of multiplicativeweights. This multiplication by complex weights can be expressed in amatrix form for both code nulling—a spectral concept—and null steering—aspatial concept. Thus, the calculations performed in the spectral codedomain correspond formally to the calculations performed in the spatialdomain. Consequently, null steering can be performed in a system usingcode-nulling simply by adding an extra dimension to the matrices usedfor calculating the complex weights and multiplying the signals by theseweights.

FIG. 10 generally depicts how weights calculated in both the code andspatial domain are used to maximize the SINR. It should be noted thatFIG. 10 is primarily a conceptual representation and is not meant toconvey the actual processing steps that occur in the method ofmaximizing SINR. As shown in FIG. 10, a three dimensional graph plotsthe relationship among code, space, and SINR. Specifically, the code andspatial domains are shown in one plane, while the SINR is plottedperpendicular to the plane defined by the code and spatial domains. TheSINR is plotted on a scale of 0 to 1 where a value of 0 indicates thatthe signal consists entirely of noise and interference while a value of1 indicates that the signal consists entirely of the signal of interest.

The code domain axis of the graph represents the various weightingvalues that can be applied to each of the tones, while the spatialdomain axis of the graph represents the weighting values that can beapplied to each of the antenna elements. As can be seen from the plot ofFIG. 10, certain weights applied in the correct combination of code andspatial values result in SINR values near 1 so that optimal signaldetection is achieved by calculating the code and spatial weights thatconverge to the “peaks” depicted in FIG. 10. The method of altering thecode and spatial domain weights so that convergence to the peak SINR isachieved is described in greater detail below with reference to themethod of maximizing SINR section. The invention combines spatial andspectral spreading and despreading to optimally remove interference fromthe received signals.

Returning to the null steering procedure that forms a portion of themethod for calculating weights in the spatial domain, the null steeringmethod, illustrated schematically in FIG. 13, provides for increaseduser capacity for each base station. As depicted in FIG. 13, a firstbeam, “beam A,” is directed by the antenna 120 using beam-formingtechniques, over a particular spatial region (i.e., the signal strengthis significant in the depicted region enclosed by solid lines). A secondbeam, “beam B,” is directed by the antenna 120 over a different spatialregion (enclosed by the dashed line in FIG. 13). Both signals includesidebands, that normally would generate interference within the adjacentsignal space, and null regions between the main beam and the sidebands.Of course, it will be appreciated that more complicated beam patternsmay be employed having several sidebands and null regions.

In accordance with one embodiment of the invention, the null regions ofbeams A and B are positioned in the direction of each of the interferingtransceivers (e.g., transceivers operating on the same tone set and/orcode as the intended transceiver). Thus, as depicted in FIG. 13, whilebeam A is directed towards remote A (since remote A is the intendedreceiver) the null of beam A is directed towards remote B (since remoteB is an interferer). Similarly, beam B is directed towards remote B(since remote B is the intended receiver) while the null of beam B isdirected towards remote-A (since remote A is an interferer). A similarweighting scheme is observed when the remotes are transmitting and thebase station is receiving. The same null-steering principle also may beapplied to reduce the interference due to neighboring base stations.

It should be noted here that multi-path reflectors may also be treatedas interfering signal sources so that null regions can be positioned tonull out signals from these reflectors. However, in one embodiment, ifthe reflectors are not significantly time varying, the reflectedinterferers are not nulled. Rather the reflected signals areadvantageously phase shifted to provide constructive interference sothat the SINR is increased.

The null resolution (i.e., the closeness in degrees of the nulls) whichthe antenna arrays are capable of providing is dependent upon severalfactors. Two main factors are the spacing of the antenna sensor elementsand the S/N ratio of the incoming signal. For instance, if the aperturesize is sufficiently large (e.g., if the sensor elements aresufficiently far apart) then a better null resolution will result. Also,if the S/N ratio of the received signal of interest is high enough, thenthe signal of interest could actually be placed partially within a null(so that some gain of the signal is lost, but the overall ratio betweenthe gain null on the interferer and the gain null on the signal ofinterest allows for effective cancellation of the interferer anddetection of the signal of interest). For example, if 15 dB of gain isnecessary to close the link for a given channel, and the S/N ratio ofthe signal of interest is 30 dB, while the S/N ratio of the interfereris 60 dB, then if a null of −70 dB is placed on the interferer, whilethe signal of interest is in the same null at about −15 dB, then theinterferer will have a net −10 dB gain and the signal of interest willhave a net 15 dB gain so that the interferer is canceled and the link isclosed. Thus, a higher S/N ratio allows the nulls to be placed closer tothe signals of interest so that a higher null resolution is achieved. Itshould be noted here that, in accordance with one advantageousembodiment of the invention, the depth of a given null is proportionalto the strength of the interferer that is to be canceled. In addition,due to the frequency diversity provided by the system, nulls can bepositioned relatively close to each other if the steering vectors(associated with the code weights) of two interfering remotes aresufficiently distinct to provide the necessary processing gain to closethe communications link.

In an alternate embodiment, the remote terminals also includedirectional antennas in one preferred embodiment so that the remoteterminals are also capable of null steering. FIG. 16 is a graph plottingantenna gain (measured in decibels) versus direction (measured indegrees). A number of base stations are represented in FIG. 16 bycrosses, while other remotes (having non-orthogonal codes) arerepresented by small circles.

In the worst case scenario, the remote is located equidistant from threebase stations (i.e., on a vertex of a hexagonal spatial cell). This caseis represented in FIG. 16 by the presence of three crosses that transmitwith substantially equivalent signal strength. These base stations areshown at approximately 0, 90°, and −90° from the zero direction of theremote antenna.

Normally, each of the base stations would be received at the same level(i.e., at −85 dB) so that substantial interference would result betweenthe three base stations when received at the remote. However, due to thebeam forming weights applied by the directional antenna of the remote,the interfering base stations (i.e., the stations at ±90°) areattenuated by approximately 50 dB (i.e., 120 dB minus 70 dB) relative tothe intended base station (i.e., the base station at 0°). Thus, due tothe fact that the beam from the receiving remote antenna is formed tohave maximum gain at the intended base station, and to have minimum gain(nulls) at the strongest interfering base stations, the remote terminalsare able to more easily discriminate between the signal of interest andinterfering signals. That is, by means of beam forming and null-steeringemployed at the remote terminals a much higher signal-to-interferenceplus noise ratio (SINR) can be obtained in much the same manner as withthe base stations.

It should be noted here that the remote terminals may also employ codenulling. In an alternate embodiment, initial code nulling weights arecalculated within the base station and transmitted to the remoteterminals. The remote terminals subsequently adapt the transmittedweights to maximize the SINR as required by the particular interferenceenvironment of each remote. By calculating the initial weights andsending these to the remote terminals, much of the intensivecalculations need not be performed within the remotes. Thus, the remoteterminals can be made more cost effectively.

In one aspect of the invention—referred to as “retrodirectivity”—thebase stations adapt the spreading and despreading weights used withinthe base stations for transmitting and receiving signals in order tomaximize the overall SINR within the communications network 100. In analternate embodiment, this may be performed, for example, by monitoringthe average bit error rate (BER) throughout the communication network100 and modifying the spreading weights at each of the base stations, aswell as each of the remote terminals, to decrease the BER.

Despread Weight Adaptation Algorithm

In one embodiment of the present invention, during the trafficestablishment phase, a series of pilot tones having known amplitudes andphases, are transmitted over the entire frequency spectrum. The pilottones are at a known level (e.g., 0 dB), and are spaced apart byapproximately 30 KHz to provide an accurate representation of thechannel response (i.e., the amplitude and phase distortion introduced bythe communication channel characteristics) over the entire transmissionband. To compensate for the channel distortion, a complex inverse(having an amplitude component and a phase component) of the channelresponse is calculated and multiplied by the incoming signals. Thisinitializes the weights during the traffic establishment phase.

In certain cases, where the channel induced fade is too deep to providean adequate signal-to-noise ratio, the tone clusters where these deepnulls occur are excised (i.e., discarded so as to not factor into thesignal during despreading).

Since the channel response varies over time, the set of complexconjugate compensation weights are periodically recalculated to insurean accurate channel estimation.

Another method of channel equalization involves equalizing the channeleffects (due, for example, to noise and known interferers) by datadirected methods. That is, rather than transmitting a known trainingsignal (such as a set of pilot tones), weights are applied to thereceived signal so as to detect a selected property of the data signal.For example, if a PSK modulation technique is used on the data, aconstant power modulus is expected in the received signal. Alternately,in a QAM signal, the data will be detected in an amplitude-phase signalconstellation plane to have substantially concentric rings. Thus, if thechannel is equalized in such a manner as to obtain the desired signalcharacteristics, there is a high probability that the transmittedsymbols will be accurately decoded at the receiver. This generaltechniques is referred to as a property restoral technique. In oneembodiment of the invention the property that is restored is the finitealphabet of the QAM or M-PSK symbol.

Of course, it will be appreciated by those skilled in the art thatalthough the channel equalization method used in accordance with theinvention is conceptually separable from other signal weighting anddecoding methods of the present invention (discussed below), the channelequalization method may implicitly include multiple cancellation anddespreading methods. Therefore, the adaptive channel equalization methodof the present invention used to maximize the SINR should not beconsidered as a separate method from the additional methods describedbelow that refer to interference cancellation and signal despreading anddecoding methods. Rather, the adaptive channel equalization method ofthe present invention should be understood to encompass a plurality ofthe below described methods.

Reciprocity and Retrodirectivity

TDD is particularly advantageous in the practice of this invention sincewith the use of TDD the linear weighting coefficients used to compensatefor channel interference during transmission and reception of theencoded signals need not be re-calculated within a station. The shorttime duration between transmission and reception by the base station,the fact that the transmission and reception occurs in the samefrequency band and only slightly separated in time (TDD), and the factthat the remote access terminals are stationary with respect to the basestations assures that the channel is approximately reciprocal. That is,the properties of the air channel between the base and the remoteterminals (i.e., those properties that introduce distortion in thetransmitted signal) are substantially the same for both reception andtransmission. Thus, substantially the same weights can be used at astation for both despreading a signal at reception and for spreading asignal at transmission. In accordance with this retrodirectivityprinciple, the base station can perform most of the computation fortransmission spreading weights when it computes the despreading weightson reception. The transmission spreading weights are merely scalarmultiples of the reception despreading weights. Similarly, in accordancewith this retrodirectivity principle, the remote station can performmost of the computation for its transmission spreading weights when itcomputes its despreading weights on reception.

In an alternate embodiment of the invention, the base station cantransmit the weights to the remote stations to be used in the nextreception at the remote station. In this manner, processing is reducedwithin the remote stations since a large portion of the intensivecalculations are performed solely within the base station. Thus, insteadof being prohibitively sophisticated, the remote terminals can be madeat a suitable size and at a reasonable expense.

Because each remote terminal stands in a different spatial relation tothe other remotes and bases within the communications network, eachremote terminal advantageously uses equalization weights that areindividually set to maximize the SINR of signals transmitted to andreceived from the base station to which the remote is assigned. This maybe accomplished in different ways. For instance, the base station maypre-emphasize the signals sent to the remote by a calculated set ofweights. Since the pre-emphasis approximately compensates for channeldistortion, the remote need not perform weight adjustment calculationsthat are as intensive as those calculated by the base station. Thus, theremotes need not include prohibitively sophisticated processingcircuitry to implement this feature of the invention.

In one aspect of the invention optimum transmit weights are calculatedbased on the signals received at the base station. This is calledretrodirectivity. When retrodirective adaptive equalization is used todetermine the set of weights used in both reception and transmission,network-wide retrodirective adaptive equalization is accomplished. Thus,the channel characteristics throughout the entire system are accountedfor in accordance with this aspect of the present invention.

Of course, as with other aspects of the invention, it will beappreciated that the reciprocity and system-wide retrodirective aspectsof the present invention may also have application in a mobileenvironment. Specifically, if the time duration between transmission andreception in the TDD system is made sufficiently small, the channel mayalso be reciprocal for mobile transceivers so that the same principlesset forth above apply in the mobile environment.

Zone Control

In a particularly preferred embodiment of the invention, a zonecontroller could be used to minimize the risk of interference betweenremote terminals that are near one another in adjacent spatial cells.According to this aspect of the invention, the zone controller isinformed of the locations of each of the remotes and base stationswithin an assigned zone. Those remote terminals that are likely tointerfere are assigned different codes and tone sets to minimize therisk of interference.

Bandwidth-on-Demand

In accordance with one aspect of the present invention, bi-directionalcommunication is established between multiple remote user units and atelephone network via the high-bandwidth base station on a user-by-userbasis. Each remote user unit upon activation, initiates communicationwith the high-bandwidth base station by indicating to one of the remoteterminals, included within the remote user unit, the amount of bandwidthdesired by the remote user unit. The remote terminals communicate withthe base station via a control channel through the air (i.e., theairlink). The high-bandwidth base station then sends informationconcerning the requested bandwidth to a central bandwidth controller,shown in FIG. 17, that determines whether or not the requested bandwidthcan be allocated to the requesting remote user unit. In this manner,bandwidth is dynamically allocated based upon the type of user unit andthe kind of data that is to be transmitted. As indicated above varyingamounts of bandwidth may be assigned by allocating additional tone setsto the requesting user.

III. A SPECIFIC EMBODIMENT OF THE INVENTION

The following description is a specific embodiment of the invention thatincludes many aspects of the description provided above. However, itshould not be interpreted to limited the scope of the invention in anyway

Frequency Definitions

The total bandwidth allocation for the airlink of this specificembodiment of the invention is 10 MHz in the range of 1850 to 1990 MHz.The total bandwidth is divided into two 5 MHz bands called the lower RFband and the upper RF band. The separation between the lowest frequencyin the lower RF band and the lowest frequency in the upper RF band (DF)is 80 MHz. The base frequency (f_(base)) for this embodiment is definedas the lowest frequency of the lower RF band. FIG. 18 shows the possibleoperational bands for this embodiment.

The lower and upper RF bands are further subdivided into sub-bands asshown in FIG. 19. The first and last 0.5 MHz of each RF band aredesignated as guard bands and are hence unused. The remaining 4 MHz ineach RF band is subdivided into four sub-bands sequentially numberedfrom 0 to 3. Furthermore, the suffix “A” indicates a sub-band within thelower RF band and “B” indicates a sub-band within the upper RF band. Thesub-bands are paired with each sub-band pair containing one sub-bandfrom the lower RF band and another from the upper RF band.

There are a total of 2560 tones (carriers) equally spaced in the 8 MHzof available bandwidth. There are 1280 tones in each band. The spacingbetween the tones (Df) is thus MHz divided by 1280, or 3.125 kHz.

The total set of tones are numbered consecutively form 0 to 2559starting from the lowest frequency tone. T_(i) is the frequency of theith tone:

T _(i) =f _(base) +f _(guard) +Df/2+(i)(Df)

for 0≦i≦1279

T _(i) =f _(base) +DF+f _(guard) +Df/2+(i)(Df)

for 1280≦i≦2559

where f_(base) is the base frequency defined in Table 2.3, f_(guard) is0.5 MHz, Df is 3.125 kHz, and DF is 80 MHz. Equivalently, therelationship may be expressed as:

T _(i) =f _(base)+500+(i+1/2)(3.125 kHz)

for 0≦i≦1279

 T _(i) =f _(base)+80500+(i+1/2)(3.125 kHz)

for 1280≦i≦2559

Each sub-band pair contains 640 tones (320 frequencies in the lowerband, and 320 in the upper band). The mapping of tones to each sub-bandis shown in FIG. 20. The set of 2560 tones is the tone space. The tonesin the tone space are used to transmit two types of data: traffic dataand overhead data. The tones used for transmission of traffic are thetraffic tones, and the rest are the overhead tones.

In an alternate embodiment of the invention, the tones can bedistributed over three or four sub-bands that are separated by largefrequency gaps, to increase immunity from interference or fading thatmay occur in any one of the sub-bands.

Traffic Tones

The traffic tones are divided into 32 traffic partitions denoted by P₀to P₃₁. (In this embodiment a traffic channel requires at least onetraffic partition.) Each traffic partition contains 72 tones as shown inFIG. 21. Tone mapping into the ith traffic partition (P_(i)) is shown inTable 2.5.

Overhead Tones

The overhead tones are used for the following channels

Forward Channels

The Common Link Channel (CLC) used by the base to transmit controlinformation to the Remote Units;

The Broadcast Channel (BRC) used to transmit broadcast information fromthe Base to all Remote Units; and

The Remote Unit Synchronization Channel (RSC) used by the base, forexample, to transmit pilot signals, frame synchronization information.

Reverse Channels

The Common Access Channels (CACs) is used to transmit messages from theRemote Unit to the Base; and

The Delay Compensation Channel (DCC) used to adjust a Remote Units TDDtiming.

For each sub-band pair, there is one grouping of tones assigned to eachchannel. These groups of tones are referred to by the name of theirchannels and their sub-band pair index (0, 1, 2, or 3). For instance,the CLC channel in sub-band pair 2 is denoted by CLC₂.

There are two different CACs in each sub-band pair: CAC_(i,0), andCAC_(i,1), where i is the sub-band pair index. The two channels may beused as either solicited (SCAC) or unsolicited (UCAC). The allocation oftones to each of these channels for the ith sub-band pair is shown inFIG. 23. Indices are provided for all tones within a given channel. Theabsolute tone index within the tone space can be determined by therelationships shown in FIG. 23. For instance:

For the forward channel, the 13th tone in the CLC channel in sub-bandpair 2, is denoted by CLC₂(13) and its absolute tone index is:

CLC ₂(13)=T _(320.2+1460) =T ₂₁₀₀

For the reverse channel, the 13th tone in the first CAC channel insub-band pair 2, is denoted by CAC_(2,0)(13) and its absolute tone indexis the same as above. FIG. 24 provides a pictorial representation of thedivision of the tone spaces into different tone groupings

Time Definitions

TDD is used by Base and the Remote Unit to transmit data and controlinformation in both directions over the same frequency channel.Transmission from the Base to the Remote Unit are called forwardtransmissions, and from the Remote Unit to the Base are called reversetransmissions.

As shown in FIG. 25, the duration of a forward transmission isT_(forward), and the duration of a reverse transmission is T_(reverse).The time between recurrent transmissions from either the Remote Unit orthe Base is TTD, the TDD period. A guard period of duration T_(f-guard)is inserted between the forward and reverse transmissions, and a guardperiod of duration T_(r-guard) is inserted between the reverse andforward transmissions.

As shown in FIG. 26, in every TDD period, there are four consecutivetransmission bursts in each direction. Data is transmitted in each burstusing multiple tones. The burst duration is T_(burst). A guard period ofduration T_(b-guard) is inserted between each burst. FIG. 27 shows thevalues of the TDD parameters.

In addition to synchronizing and conforming to the TDD structure definedin the last section, both the Base and the Remote Unit must synchronizeto the framing structure. The framing structure is shown in FIG. 28. Thesmallest unit of time shown in this figure is a TDD period. Two TDDperiods make a subframe, eight subframes make a frame, and 32 framesmake a superframe.

Frame synchronization is performed at the superframe level. The frameand subframe boundaries are determined from the superframe boundary.

In this embodiment we could potentially reuse all available frequenciesin every spatial cell. However, initially a reuse factor of 2 is used.Each Remote Unit is assigned to a Sub-band Pair depending on itslocation within the spatial cell and the traffic loading of the Sub-bandPair. As shown in FIG. 29 each Remote Unit may be assigned two of thefour sub-band pairs depending on its location. For example, an RemoteUnit in the north-eastern part of the spatial cell in FIG. 29 can beassigned Sub-band Pair 0 or Sub-band Pair 2. Of course this reusestrategy reduces capacity to half of the maximum potential capacity. Thesame Sub-band Pair assignment is used in all spatial cells as shown inFIG. 30.

Forward Channel Format

The physical layer has three possible implementations based on thedesired range (or quality) of transmission. The physical layer managesthe trade-offs between bandwidth efficiency (bits/symbol) andtransmission coverage by providing three modes of operation:

High capacity mode (short range): 3 bits/symbol

Medium capacity mode (medium range): 2 bits/symbol

Low capacity mode (long range): 1 bits/symbol

Each mode employs different details in the coded modulation scheme and,hence, somewhat different formats. Nevertheless, there is abundantsymmetry, redundancy, and common elements for the three modes.

High Capacity Mode

In high capacity mode, one traffic partition is used in one trafficchannel. In medium and low capacity modes, two and three trafficpartitions are used, respectively. The Base transmits information tomultiple Remote Units in its spatial cell. This section describes thetransmission formats for a 64 kbits/sec traffic channel, together with a4 kbps Link Control Channel (LCC) from the Base to a single Remote Unit.The block diagram for the upper physical layer of the Base transmitterfor high capacity mode is shown in FIG. 31, which shows data processingfor one forward channel burst. (The boundary between the upper and thelower physical layers is where the baseband signals are translated intofrequency tones. The lower physical layer can then be regarded as thecommon element of the various modes and directions of transmission.) Thelarge shaded area shows the processing required for one traffic channelat the Base. The remainder of the diagram shows how various trafficchannels are combined. The details of each block in the diagram arediscussed throughout this section.

The binary source delivers data to the Base transmitter at 64 kbits/sec.This translates to 48 bits in one forward transmission burst.

The information bits are encrypted according to the triple dataencryption standard (DES) algorithm.

The encrypted bits are then randomized in the data randomization block.The bit to octal conversion block converts the randomized binarysequence into a sequence of 3-bit symbols. The symbol sequence isconverted into 16 symbol vectors. (In this description, the term vectorgenerally refers to a column vector. A vector is generally complexunless otherwise stated. Generally, column vectors are denoted by boldlower-case characters, while row vectors are denoted by the samecharacters with a transpose operation, denoted by a superscript T.Another widely used vector form used here is a conjugate transposevector referred to here as Hermetian.) One symbol from the LCC is addedto form a vector of 17 symbols.

The 17-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the LCCsymbol). This process employs convolutional encoding that converts theinput symbol (an integer between 0 and 7) to another symbol (between 0and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is signal withinthe set of 16 QAM (or 16PSK) constellation signals. (The term signalwill generally refer to a signal constellation point.)

A link maintenance pilot signal (LMP) is added to form an 18-signalvector, with the LMP as the first elements of the vector. The resulting(18×1) vector d_(fwd) is pre-multiplied by a (18×18) forward smearingmatrix C_(fwd-smear) to yield a (18×1) vector b.

Vector b is element-wise multiplied by the (18×1) gain preemphasisvector g_(fwd)(p) to yield another (18×1) vector, c, where p denotes thetraffic channel index and is an integer in the range [0, M_(base)] whereM is the maximum number of traffic channels that can simultaneously becarried over one traffic partition. Vector c is post-multiplied by a(1×32) forward spatial and spectral spreading vector g^(H) _(fwd)(p) toyield a (18×32) matrix R(p). The number 32 results from multiplying thespectral spreading factor 4 and spatial spreading factor 8. The 18×32matrices corresponding to all traffic channels carried (on the sametraffic partition) are then combined (added) to produce the resulting18×32 matrix S_(fwd).

The matrix S_(fwd) is partitioned (by groups of four columns) into eight(18×4) submatrices (A₀ to A₇). (The indices 0 to 7, corresponds to theantenna elements over which these symbols will eventually betransmitted.) Each submatrix is mapped to tones within one trafficpartition (denoted by partition A in FIG. 31) according to the mappingdiscussed in FIG. 22 and sent to the lower physical layer.

The lower physical layer places the baseband signals in discrete Fouriertransfer (DFT) frequency bins where the data is converted into the timedomain and sent to its corresponding antenna elements (0 to 7) fortransmission over the air. The details of the lower physical layer arediscussed below.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next forward transmission burst. Thevarious steps in the transformation of binary data are shown in FIG. 32.To keep the diagram simple, the spreading and traffic channel combinerfunctions are shown in one step.

Medium Capacity Mode

The block diagram for the upper physical layer of the Base transmitterfor the medium capacity mode is shown in FIG. 33. The primary differencebetween the transmission formats for high and medium capacity modes isthe use of different trellis encoding schemes. In medium capacity mode,an 8QAM (or 8PSK) rate 2/3 trellis encoder (compared to a 16QAM or 16PSKrate 3/4 bits in one forward transmission burst, two traffic partitions(A and B) are used.

The binary source delivers binary data to the Base transmitter at 64kbits/sec. For one forward channel burst, this translates to 48 bits.The information bits are encrypted according to the triple DESalgorithm. The encrypted bits are then randomized in the datarandomization block. The bit to-bit conversion block converts therandomized binary sequence into a sequence of 2-bit symbols. The symbolsequence is converted into 24 symbol vectors. Two symbols from the LCCare added, and eight ones are inserted at the end of the sequence toform a vector of 34 symbols. (The two symbols for the LCC carry onlythree bits of LCC information. The least significant bit, LSB, of thesecond LCC symbol is always set to one.)

The 34-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the secondLCC symbol). This process employs convolutional encoding that convertsthe input symbol (an integer between 0 and 3) to another symbol (between0 and 7) and maps the encoded symbol to its corresponding 8 QAM (or8PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 34 elements where each element is a signal withina set of 8QAM (or 8PSK) constellation signals.

The 34-element vector is divided into two 17-element vectors. An LMP isadded to each of the vectors to form two 18-element vectors d_(fwd) andd′_(fwd), where the LMP is the first element of these vectors. Eachresulting vector is pre-multiplied by a (18×18) forward smearing matrixC_(fwd-smear) to yield another two (18×1) vector b and b′. Vectors b andb′ are then element-wise multiplied by two (18×1) gain preemphasisvectors g_(fwd)(p) and g′_(fwd)(p) to yield two (18×1) vectors c and c′,where p denotes the traffic channel index. Each vector ispost-multiplied by its corresponding (1×32) Forward Spatial and Spectralspreading Vector (g^(H) _(fwd)(p) or (g′^(H) _(fwd)(p)) to yield two(18×32 matrices R(p) and R′(p).

The various 18×32 matrices corresponding to all traffic channels carriedon traffic partition A are combined to produce the 18×32 matrix S_(fwd).Similarly, matrices from those traffic channels carried on TrafficPartition B are combined to produce the 18×32 matrix S′_(fwd).

Matrix S_(fwd) is partitioned (by groups of four columns) into eight(18×4) submatrices (A₀ to A₇). Each submatrix is mapped into toneswithin partition A according to the mapping discussed in FIG. 22 and issent to the lower physical layer. Similarly, matrix S′_(fwd) ispartitioned into eight (18×4) submatrices (A′₀ to A′₇). Each submatrixis mapped into tones with partition B according to the mapping discussedin FIG. 22 and is sent to the lower physical layer.

The lower physical layer places the baseband signals in DFT frequencybins where the data is converted into the time domain and sent to itscorresponding antenna element (0 to 7) for transmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next forward channel transmission burst.The various steps in the transformation of binary data are shown in FIG.34. To keep the diagram simple, the spreading and traffic channelcombiner functions are shown in one step. The block diagram for theupper physical layer of the Base transmitter for low capacity mode isshown in FIG. 35.

Low Capacity Mode

The primary difference between the transmission formats for high and lowcapacity modes is the use of different trellis encoding schemes. In lowcapacity mode, a rate 1/2 trellis encoder (compared to a rate 3/4encoder for high capacity mode) is employed. To transmit 48 bits in oneforward transmission burst, three Traffic Partitions (A, B, and C) areused.

The binary source delivers binary data to the Base transmitter at 64kbits/sec. For one forward channel burst, this translates to 48 bits.The information bits are encrypted according to the Triple DESalgorithm. The encrypted bits are then randomized in the datarandomization block. The 48 bits are formed into a vector. Three symbolsfrom the LCC are then added to form a vector of 51 symbols. The51-symbol vector is trellis encoded. The trellis encoding starts withthe most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the thirdLCC symbol). This process employs convolutional encoding that convertsthe binary input symbol (0 or 1) to another symbol (0, 1, 2, or 3) andmaps the encoded symbol to its corresponding QPSK signal constellationpoint. The output of the trellis encoder is therefore a vector of 51elements where each element is a signal within the set of QPSKconstellation signals.

The 51-element vector is divided into three 17-element vectors. An LMPis added to each of the vectors to form three 18-element vectorsd_(fwd), d′_(fwd), and d″_(fwd), where the LMP is the first element ofthese vectors. Each resulting vector is pre-multiplied by a (18×18)forward smearing matrix C_(fwd-smear) to yield another three (18×1)vectors b, b′, and b″. Vectors b, b′, and b″ are then element-wisemultiplied by their respective (18×1) gain preemphasis vectorsg_(fwd)(p), g′_(fwd)(p), and g″ to yield three (18×1) vectors c, c′, andc″, where p denotes the traffic channel index. Each vector ispost-multiplied by its corresponding (1×32) forward spatial and spectralspreading vector (g^(H) _(fwd)(p), g′^(H) _(fwd)(p), or g″^(H)_(fwd)(p)) to yield three (18×32) matrices R(p), R′(p), and R″(p).

The various 18×32 matrices corresponding to traffic channels carried ontraffic partition A are combined to produce the 18×32 matrix S_(fwd).Similarly, matrices from those traffic channels carried on trafficpartitions B and C are combined to produce two 18×32 matrices, S′_(fwd)and S″_(fwd), respectively. Matrix S_(fwd) is partitioned (by groups offour columns) into eight (18×4) submatrices (A₀ to A₇). Each submatrixis mapped into tones within partition A according to the mappingdiscussed in FIG. 22 and is sent to the lower physical layer. MatrixS′_(fwd) is partitioned into eight (18×4) submatrices (A′₀ to A′₇). Eachsubmatrix is mapped into tones within partition B and is sent to thelower physical layer. Similarly, matrix S″_(fwd) is partitioned intoeight (18×4) submatrices (A″₀ to A″₇). Each submatrix is mapped intotones within partition C and is sent to the lower physical layer. Thelower physical layer places the baseband signals in DFT frequency binswhere the data is converted into the time domain and sent to itscorresponding antenna element (0 to 7) for transmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next forward channel transmission burst.The various steps in the transformation of binary data are shown in FIG.36. To keep the diagram simple, the spreading and traffic channelcombiner functions are shown in one step. Similarly, the encryption andrandomization functions are also shown in one step.

Encryption/Decryption

The 64 kbps binary source delivers bits to the encryption module 48 bitsat a time. The encryption function is a three-stage cascade of the DESalgorithm as shown in FIG. 37.

Trellis Encoding/Decoding

The trellis encoding technique consists of convolutional encodingfollowed by a signal mapping. The three modes of the physical layer usedifferent trellis codes. For high capacity mode, there are two possiblesignal constellations: 16PSK and 16QAM.

The rate 3/4 convolutional encoder for the 16 PSK constellation is shownin FIG. 38. The convolutional encoder employs an 8-state (k=4)¹⁴ rate1/2 mother encoder that encodes one bit out of a 3-bit input symbol, andpasses the remaining bits uncoded.

The rate 1/2 convolutional encoder for the 16PSK constellation may bedescribed by the generator polynomials (G₀=04, G₁=13), in octalrepresentation. Equivalently, the polynomial representation is:

G ₀ =D

G ₁ =D ³ +D ²+1

The rate 3/4 convolutional encoder for the 16QAM constellation is shownin FIG. 39. The convolutional encoder employs an 8-state (k=4) rate 1/2mother encoder that encodes one bit out of a 3-bit input symbol, andpasses the remaining bits uncoded.

The rate 1/2 convolutional encoder for the 16QAM constellation may bedescribed by the generator polynomials (G₀=17, G₁=13), in octalrepresentation. Equivalently, the polynomial representation is:

G ₀ =D ³ +D ² +D+1

G ₁ =D ³ +D ²1

The two highest-order bits of the input symbol (x₂,x₁) are passedthrough uncoded to form the two highest-order bits of the output symbol(y₃,y₂). The lowest-order bits of the input symbol (x₀) enters the rate1/2 mother encoder (shown as the shaded box) to produce two lowest-orderbits of the output symbol (y₁,y₀).

The next step in the trellis encoding process is to map the outputsymbol onto a signal in the 16QAM (or 16PSK) constellation. Theparticular mappings for the 16QAM and 16PSK constellations are shown inFIG. 40.

The resulting trellis encoder output is one of 16 possible complexnumbers within the 16QAM (or 16PSK) constellation shown in FIG. 40. Theactual value of each constellation point (signal) is shown in FIG. 41.The points on the constellation have been chosen so that the averageenergy of the signal is 1.

In medium capacity mode, a rate 2/3 trellis code with either 8QAM or8PSK signal mapping is employed. The convolutional encoder for the 8PSKconstellation is shown in FIG. 42. It employs a 32 state (k=6) rate 1/2mother encoder that encodes one bit out of a 2-bit input symbol, andpasses the remaining bit uncoded. The rate 1/2 convolutional encoder maybe described by the generator polynomials (G₀=10, G₁=45), in octalrepresentation. Equivalently, the polynomial representation is:

G ₀ =D ²

 G ₁ =D ⁵ +D ³ +D ²+1

The convolutional encoder for the 8QAM constellation is shown in FIG.43. It employs a 32-state (k=6) rate 1/2 mother encoder that encodes onebit out of a 2-bit input symbol, and passes the remaining bit uncoded.The rate 1/2 convolutional encoder may be described by the generatorpolynomials (G₀=53, G₁=75), in octal representation. Equivalently, thepolynomial representation is:

G ₀ =D ⁵ +D ⁴ +D ²+1

G ₁ =D ⁵ +D ³ +D ²+1

The highest-order bit of the input symbol (x₁) is passed through uncodedto form the highest-order bit of the output symbol (y₂). Thelowest-order bit of the input symbol (x₀) enters the rate 1/2 motherencoder to produce the two lowest-order bits of the output symbol(y₁,y0).

The next step in the trellis encoding process is to map the outputsymbol onto a signal in the 8QAM (or 8PSK) constellation. The particularmappings for the 8QAM and 8PSK constellations are shown in FIG. 44. Theresulting trellis encoder output is one of eight possible complexnumbers within the 8QAM (or 8PSK) constellation shown in FIG. 44. Theactual value of each constellation point (signal) is shown in FIG. 45.The points of the constellation have been chosen so that the averageenergy of the signal is 1.

In low capacity mode, the de facto standard rate 1/2 encoder, shown inFIG. 46, together with QPSK mapping is employed. The only bit of theinput symbol (x₀) enters the rate 1/2 mother encoder to produce the twobits of the output symbol (y₁,y₀).

The next step in the trellis encoding process is to map the outputsymbol onto a signal in the QPSK constellation. The particular mappingfor the QPSK constellation is shown in FIG. 46 referred to by naturalmapping. The resulting trellis encoder output is one of four possiblecomplex numbers within the QPSK constellation shown in FIG. 47. Theactual value of each constellation point (signal) is shown in FIG. 48.The points on the constellation have been chosen so that the averageenergy of the signal is one.

Cluster Smearing/Desmearing

This section defines the smearing matrix C_(fwd-smear). The input to thesmearing block is the (18×1) vector d_(fwd). The output of the smearingoperation (vector b) can then be described by the matrix multiplicationof d_(fwd) and the (18×18) smearing matrix C_(fwd-smear). That is

b=C _(fwd-smear) d _(fwd)

C_(fwd-smear) is the constant valued matrix shown below:

1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 α ^(βδ0) 0 0 0 0 0 0 0 0 0 0 0 0 0 00 0 α 0 ^(βδ1) 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 α 0 0 ^(βδ2) 0 0 0 0 0 0 00 0 0 0 0 0 0 α 0 0 0 ^(βδ3) 0 0 0 0 0 0 0 0 0 0 0 0 0 α 0 0 0 0 ^(βδ4)0 0 0 0 0 0 0 0 0 0 0 0 α 0 0 0 0 0 ^(βδ5) 0 0 0 0 0 0 0 0 0 0 0 α 0 0 00 0 0 ^(βδ6) 0 0 0 0 0 0 0 0 0 0 α 0 0 0 0 0 0 0 ^(βδ7) 0 0 0 0 0 0 0 00 α 0 0 0 0 0 0 0 0 ^(βδ8) 0 0 0 0 0 0 0 0 α 0 0 0 0 0 0 0 0 0 ^(βδ9) 00 0 0 0 0 0 α 0 0 0 0 0 0 0 0 0 0 ^(βδ10) 0 0 0 0 0 0 α 0 0 0 0 0 0 0 00 0 0 ^(βδ11) 0 0 0 0 0 α 0 0 0 0 0 0 0 0 0 0 0 0 ^(βδ12) 0 0 0 0 α 0 00 0 0 0 0 0 0 0 0 0 0 ^(βδ13) 0 0 0 α 0 0 0 0 0 0 0 0 0 0 0 0 0 0^(βδ14) 0 0 α 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 ^(βδ15) 0 α 0 0 0 0 0 0 0 00 0 0 0 0 0 0 0 ^(βδ16)

where,

a=(r _(LMP)/(1+r _(LMP)))^(1/2)

b=(1/(1+r _(LMP)))^(1/2)

and r_(LMP) is the ratio of pilot to data power that is a physical layerprovisionable parameter whose value is nominally set to one.

Gain Preemphasis

This section discusses the gain preemphasis matrix g_(fwd)(p) shown inFIG. 31. The input to the gain preemphasis block is the (18×1) vector b.The output of the gain preemphasis operation (Vector c) is theelement-wise multiplication of vector b and the gain preemphasis vectorg_(fwd)(p):

c=b·g _(fwd)(p)

where·represents element-wise vector multiplication. The elements ofg_(fwd)(p) are derived using information received at the Base. Thederivation of these weights are implementation dependent.

Spectral and Spatial Spreading

This section defines the (1×32) forward spatial and spectral spreadingvector g^(H) _(fwd)(p) shown in FIG. 31. The input to the spectral andspatial spreading block is the 18-element vector c. The output of thespectral and spatial spreading operation, (18×32) matrix R(p), is thematrix multiplication of c and the (1×32) spectral and spatial spreadingvector g^(H) _(fwd)(p):

R(p)=cg ^(H) _(fwd)(p)

where,

g ^(H) _(fwd)(p)=[g ₀ g ₁ g ₂ . . . g ₃₀ g ₃₁]

The elements of vector g^(H) _(fwd)(p) are transmit spreading weightscalculated throughout the transmission. The algorithm for the derivationof these weights is implementation dependent. However, to clarify theprocedure, a specific algorithm for the derivation of these weights isdescribed below.

The Base derives its new weights based on the most recent data receivedon the reverse channel. The transmit weights are a scaled version of thereceived weights using eight antenna inputs with four receivefrequencies per antenna. The receive weight vector w^(H) _(rev)(p) has32 elements (w₀−w₃₁) that are mapped to spatial and spectral componentsas shown in FIG. 49.

For the Base traffic establishment procedure, the transmit weights(g₀−g₃₁) are calculated according to the following equation:

g ^(H) _(fwd)(p)=a _(fwd)(n)h(k _(fwd) w ^(H) _(rev)(p))

where k_(fwd) is the Base transmission constant, a_(fwd)(n) is the Basegain ramp-up factor for the nth packet, and h(.) is a function thatlimits the norm of its argument to 23 dBm

h(v)=v

for ∥v∥²<23 dBm

h(v)=23 dBm (scale factor) (v/∥v∥²)

otherwise

For the Base steady-state procedure, receive weights are adaptivelycalculated using the following equation:

w _(rev)(p)=R ⁻¹ _(xx) r _(xy)

where

w_(rev)(p) is the (32×1) weight vector;

r_(xy) is an estimate of the (32×1) cross-correlation vector of thereceived (32×1) vector x and the despread data y, multiplied by anestimate of the channel equalization weights; and

R⁻¹ _(xx) is an estimate of (32×32) inverted auto-correlation matrix ofthe received vector x. (R⁻¹ _(xx) may be computed using the RecursiveModified Gramm-Schmidt (RMGS) algorithm.)

r_(xy) is cross-correlated against the despread data after a resmearingstep and a gain pre-emphasis reapplication step.

The receive weights (w₀−w₃₁) are mapped to spatial and spectralcomponents according to the mapping shown in FIG. 49. The transmitweights (g₀−g₃₁) are a scaled version of the receive weights. Thescaling is made according to the following equation:

g ^(H) _(fwd)(p)=k _(fwd) w ^(H) _(rev)(1)

where k_(fwd) is the Base steady-state transmission constant.

Correlation estimates are computed over eight reverse-channel bursts.The new despreading weights are applied to eight reverse channel burstwith no delay. The spreading weights are applied to eight forwardchannel bursts after an 4-burst delay. Correlation estimates are madeusing an exponentially averaged block summation. The exponential decayconstant is provisionable with a nominal value of 0.7.

An illustrative flowchart of an embodiment of the adaptive solution ofspectral and spatial weights is shown in FIG. 85.

Forward Control Channel Transmission Format

The block diagram for the physical layer of the Common Link Channel(CLC) channel transmissions is shown in FIG. 50. A CLC message is a64-bit binary sequence. The bit to di-bit conversion block converts thebinary sequence into a sequence of 2-bit symbols of length 32. Thevector formation block converts the symbol sequence into a (32×1)vector. Each element of the resulting vector is mapped into itscorresponding signal in the QPSK signal constellation to form another(32×1) vector, s. The mapping for the QPSK signal is shown in FIG. 51.

The resulting vector is passed through two parallel paths. In the firstpath, the vector s is sent directly for spectral and spatial spreadingthat involves post-multiplying it by the (1×32) spreading vector g_(clc)^(H):

g ^(H) _(clc) =[g 0 g 1 g 2 . . . g 30 g 31]

(g^(H) _(clc) is discussed further below.) The resulting (32×32) matrixis D_(clc). Matrix D_(clc) is then sent to the antenna demultiplexerwhere it is partitioned (by groups of 4 columns)into eight (32×4)submatrices A₀ to A₇. The elements of these matrices will ultimately betransmitted on antennas 0 to 7, respectively.

In the second path, the vector s is code-gated. The code-gatingoperation is described by the element-wise multiplication of the (32×1)vector s with a (32×1) code-gating vector Y_(clc). The resulting (32×1)vector is S′:

s′=s·i _(clc)

The vector i_(clc) is described below.

The resulting (32×1) vector s′ is sent for spectral and spatialspreading that involves post-multiplying it by the (1×32) spectral andspatial spreading vector g_(clc) ^(H). The resulting (32×32) matrix isD′_(clc) Matrix D′_(clc) is then sent to the antenna demultiplexer whereit is partitioned (by groups of 4 columns) into eight (32×4) submatricesA′₀ to A′₇. Each of these matrices (A′₀ to A′₇) and (A′₀ to A′₇) is thensent to a time demultiplexer where it is further partitioned (by groupsof 4 rows) into eight (4×4) submatrices. This yields 128 (4×4) matrices(D₀ to D₆₃) and (D′₀ to D′₆₃).

The transmission of one 64 bit CLC message requires 16 forward channelbursts or 4 TDD periods. In each of these bursts, eight (one for eachantenna) of the (4×4) matrices are mapped onto tones and sent to thelower physical layer for transmission over the air. The interleaving andtone mapping functions are described herein.

The vector g_(clc) is defined as the Kronecker product of a (8×1)spatial spreading vector d and a (4×1) spectral spreading vector f:

g _(clc) =kron(d,f)

where d, is given by ${\quad \begin{matrix}d_{0} \\d_{1} \\d_{2} \\d_{3} \\d_{4} \\d_{5} \\d_{6} \\d_{7}\end{matrix}\quad }$

and f is given by ${\quad \begin{matrix}f_{0} \\f_{1} \\f_{2} \\f_{3}\end{matrix}\quad }$

The resulting vector g_(clc) is given by: ${\quad \begin{matrix}{\delta \quad 0{f0}} \\{\delta \quad 0{f1}} \\{\delta \quad 0{f2}} \\{\delta \quad 0{f3}} \\{\delta \quad 0{f0}} \\\vdots \\{\delta \quad 7{f2}} \\{\delta \quad 7{f3}}\end{matrix}\quad }$

The g_(clc) ^(H) is the conjugate transpose of g_(clc).

The spreading vector f is a column of the (4×4) Hadamard matrix H₄, thatmay be chosen randomly by the Base.

The spreading vector d is the kth column of the (8×72) CLC SpatialSpreading Weights Table. The column index k is provided by the MAC layerthrough the parameter CLC beam.

A (N×N) Hadamard matrix denoted by H_(N) is obtained by the followingrecursion: H_(2n) equals ${\quad \begin{matrix}H_{n} & H_{n} \\H_{n} & H_{n}\end{matrix}}\quad $

where H₀ is initialized at 1. For instance, the 4×4 Hadamard matrix (H₄)is: ${\quad \begin{matrix}1 & 1 & 1 & 1 \\1 & {- 1} & 1 & {- 1} \\1 & 1 & {- 1} & {- 1} \\1 & {- 1} & {- 1} & 1\end{matrix}\quad }$

The code-gating vector i_(clc) is:

i _(clc) =b _(clc) ·h _(clc)

where the vector h_(clc) is the 0th column of the (32×32) Hadamardmatrix (H₃₂), that is the all ones vector and the ith element of the(32×1) vector b_(clc) is given by:

b _(clc) =e ^(j2pik)offset^(/32)

The k_(offset) (an integer between 0 and 31) is the Base station offsetcode (BSOC) for the transmitting Base.

Interleaving

There are 16 burst in every CLC transmission (burst 0 to burst 15). Foreach antenna, the interleaver outputs one of the 16 possible (4×4)matrices in each burst. FIG. 52 shows the order of the transmission usedby the interleaver.

Tone Mapping

There are 128 (4×4) matrices to be mapped onto tones for transmissionover the air. FIG. 53 shows the mapping of a (4×4) matrix at the outputof the interleaver into tones. The absolute tone indices can be obtainedusing FIG. 23.

The Broadcast Channel

The block diagram for the physical layer of BRC channel transmissions isshown in FIG. 54. The block diagram is very similar to that for the CLCshown in FIG. 50. However, for the sake of completeness, and to pointout the small differences, the details of the BRC transmission formatare included in this section.

The primary differences between the CLC and the BRC transmissions on theforward channel are:

The Base uses all our BRC channels (in the four sub-band pairs) whilefor the CLC, channel selection is based on its operating sub-band pair

The Base forms 10 spatial beams (activated sequentially) to cover allthe RUs in one hemisphere, that means that the time to broadcast a BRCmessage is ten times as long as transmission of a CLC message

A BRC message is a 64-bit binary sequence. The bit to di-bit conversionblock converts the binary sequence into a sequence of 2-bit symbols oflength 32. The vector formation block converts the symbol sequence intoa (32×1) vector. Each element of the resulting vector is mapped into itscorresponding signal in the QPSK signal constellation to form another(32×1) vector s. The mapping for the QPSK signal is identical to thatfor the CLC shown in FIG. 51.

The resulting vector is passed through two parallel paths. In the first,path, the vector s is sent directly for spectral and spatial spreadingthat involves post-multiplying it by the (1×32) spectral and spatialspreading vector g_(brc) ^(H).

g ^(H) _(brc) =[g 0 g 1 g 2 . . . g 30 g 31]

The g_(brc) ^(H) is discussed below.

The resulting (32×32) matrix is D_(brc). Matrix D_(brc) is then sent tothe antenna demultiplexer where it is partitioned (by groups of fourcolumns) into eight (32×4) submatrices A₀ to A₇. The elements of thesematrices will ultimately be transmitted on antennas 0 to 7,respectively.

In the second path, the vector s is code-gated. Code-gating is describedby the element-wise multiplication of the (32×1) vector s with a (32×1)code-gating vector i_(brc). The resulting (32×1) vector is s′:

s′=s·Y _(brc)

The vector Y_(brc) is described below.

The resulting (32×1) vector s′ is sent for spectral and spatialspreading, that involves post-multiplying it by the (1×32) spreadingvector g_(brc) ^(H). The resulting (32×32) matrix is D′_(brc). MatrixD′_(brc) is then sent to the antenna demultiplexer where it ispartitioned (by groups of four columns) into eight (32×4) submatricesA′₀ to A′₇. Each of these matrices (A₀ to A₇) and (A′₀ to A′₇) is thensent to a time demultiplexer where they are further partitioned (bygroups of four rows) into eight (4×4) submatrices. This yields 128 (4×4)matrices (D₀ to D₆₃) and (D′₀ to D′₆₃).

For one spatial beam, the transmission of one 64-bit BRC messagerequires 16 forward channel bursts or four TDD periods. In each of thesebursts, eight (one for each antenna) of the (4×4) matrices are mappedonto tones and sent to the lower physical layer for transmission overthe air. The interleaving and tone mapping functions are describedherein.

This process is repeated 10 times to provide 10 spatial beams withdifferent directions so that all the RUs in a spatial cell can detectthe broadcast message. The details of the beam sweeping are describedbelow. The duration of a BRC transmission is therefore 160 bursts or 40TDD periods.

The vector g_(brc) is defined as the Kronecker product of a (8×1)spatial spreading vector d and a (4×1) spectral spreading vector f:

g _(brc) =kron(d,f)

where d, is given by ${\quad \begin{matrix}d_{0} \\d_{1} \\d_{2} \\d_{3} \\d_{4} \\d_{5} \\d_{6} \\d_{7}\end{matrix}\quad }$

and f is given by ${\quad \begin{matrix}f_{0} \\f_{1} \\f_{2} \\f_{3}\end{matrix}\quad }$

The resulting vector g_(brc) is given by: ${\quad \begin{matrix}{\delta \quad 0{f0}} \\{\delta \quad 0{f1}} \\{\delta \quad 0{f2}} \\{\delta \quad 0{f3}} \\{\delta \quad 0{f4}} \\\vdots \\{\delta \quad 7{f2}} \\{\delta \quad 7{f3}}\end{matrix}\quad }$

The g_(brc) ^(H) is the conjugate transpose of g_(brc). The spreadingvector f is a column of the (4×4) Hadamard matrix H₄, that may be chosenrandomly by the Base. The spreading vector d is a column of the BRCSpatial Spreading Weights Table that will be described in the nextrelease of this document. The Base transmits simultaneously on all thesub-band pairs; for each sub-band pair, 10 different spatial beams areformed and activated sequentially to cover al the RUs in the spatialcell.

The code-gating vector i_(brc) is:

i _(brc) =b _(brc) h _(brc)

where the vector h_(brc) is the 0th column of the (32×32) Hadamardmatrix (H₃₂), that is the all ones vector and the ith element of the(32×1) vector b_(clc) is given by:

b _(brc) =e ^(j2pik)offset^(/32)

The k_(offset) (an integer between 0 and 31) is the BSOC for thetransmitting Base.

For every spatial beam, there are 16 bursts in every BRC transmission(burst 0 to burst 15). For each antenna, the interleaver outputs one ofthe 16 possible (4×4) matrices in each burst. The interleaving rule isidentical to the CLC interleaving rule shown in FIG. 52. There are atotal of spatial beams. This process is therefore repeated sequentiallyten times, once for every spatial beam.

For every spatial beam, there are 128 (4×4) matrices to be mapped ontotones for transmission over the air. FIG. 55 shows the mapping of a(4×4) matrix at the output of the interleaver into tones. The absolutetone indices can be obtained using FIG. 23.

Broadcast channel signals are spatially beamformed and transmittedsequentially via ten predetermined team patterns per sub-band pair. Thisresults in four broadcast channel signals (one per sub-band pair) thatare simultaneously swept through each spatial cell. This is shown inFIG. 56.

Each BRC message requires 40 TDD intervals or 120 ms to transmit. NewBRC message may only be started on even frame boundaries. Each of thefour sub-band pairs transmits the same BRC message at the same time andthe BRC beam sweeps are synchronous within a spatial cell and for allBases within a this embodiment system. BRC beams are swept in aclockwise pattern.

Reverse Channel Format

As for the forward channel transmissions, there are three differentpossible implementations of the physical layer. We refer to these modesas:

High capacity mode (short range): 3 bits/symbol

Medium capacity mode (medium range): 2 bits/symbol

Low capacity mode (long range): 1 bit/symbol.

High Capacity Mode

The block diagram for the upper physical layer of the Remote Unittransmitter for high capacity mode is shown in FIG. 57.

The binary source delivers binary data to the Remote Unit transmitter at64 kbits sec. For one reverse channel burst, this translates to 48 bits.The information bits are encrypted according to the triple DESalgorithm. The encrypted bits are then randomized in the datarandomization block.

The bit to octal conversion block converts the randomized binarysequence into a sequence of 3-bit symbols. The symbol sequence isconverted into 16-symbol vectors. One symbol from the LCC is added toform a vector of 17 symbols.

The 17-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the LCCsymbol). This process employs convolutional encoding that converts theinput symbol (an integer between 0 and 7) to another symbol (between 0and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is a signal withinthe set of 16QAM (or 16PSK) constellation signals.

A LMP signal is added to form an 18-signal vector, with the LMP as thefirst element of this vector. The resulting vector d_(rev) ispre-multiplied by a (18×18) reverse smearing matrix C_(rev-smear) toyield a (18×1) vector b. Vector b is post-multiplied by a (1×4) reversespreading vector g^(H) _(rev) to yield a (18×4) matrix S_(rev). Elementsof matrix S_(rev) are mapped to tones within traffic partition Aaccording to the mapping discussed in FIG. 22 and are sent to the lowerphysical layer. The lower physical layer places the baseband signals intheir corresponding DFT frequency bins where the data is converted intothe time domain and sent for transmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next reverse channel transmission burst.The various steps in the transformation of binary data are shown in FIG.58.

Medium Capacity Mode

The block diagram for the upper physical layer of the Remote Unittransmitter for medium capacity mode is shown in FIG. 59.

The binary source delivers binary data to the Remote Unit transmitter at64 kbits/sec. For one reverse channel burst, this translates to 48 bits.The information bits are encrypted according to the triple DESalgorithm. The encrypted bits are then randomized in the datarandomization block. The bit to di-bit conversion block converts therandomized binary sequence into a sequence of 2-bit symbols. The symbolsequence is converted into 24 symbol vectors. Two symbols from the LCCare added, and eight ones are inserted at the end of the sequence toform a vector of 34 symbols.

The 34-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the secondLCC symbol). This process employs convolutional encoding that convertsthe input symbol (an integer between 0 and 3) to another symbol (between0 and 7) and maps the encoded symbol to its corresponding 8 QAM (or8PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 34 elements where each element is a signal withinthe set of 8QAM (or 8PSK) constellation signals.

The 34-element vector is divided into two 17-element vectors. An LMP isadded to each of the vectors to form two 18-signal vectors d_(rev) andd′_(rev), with the LMP as the first element of these vectors. Eachresulting vector is pre-multiplied by a (18×18) reverse smearing matrixC_(rev-smear) to yield another two (18×1) vectors b and b′. Each vectoris post-multiplied by its corresponding (1×4) reverse spreading vector(g^(H) _(rev) or g′^(H) _(rev)) to yield two (18×4) matrices S_(rev) andS′_(rev). Elements of matrix S_(rev) are mapped to tones within trafficpartition A according to the mapping discussed in FIG. 22 and are sentto the lower physical layer. Similarly, elements of matrix S′_(rev) aremapped to tones within traffic partition B and are sent to the lowerphysical layer. The lower physical layer places the baseband signals inDFT frequency bins where the data is converted into the time domain andsent for transmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next reverse channel transmission burst.The various steps in the transformation of binary data are shown in FIG.60.

Low Capacity Mode

The block diagram for the upper physical layer of the Remote Unittransmitter for low capacity mode is shown in FIG. 61. The binary sourcedelivers binary data to the Remote Unit transmitter at 64 kbits/sec. Forone reverse channel burst, this translates to 48 bits. The informationbits are encrypted according to the triple DES algorithm. The encryptedbits are then randomized in the data randomization block. The 48 bitsare formed into a vector, and three bits from the LCC are added to forma vector of 51 bits.

The 51-bit vector is trellis encoded. The trellis encoding starts withthe most significant bit (first element of the vector) and is continuedsequentially until the last element of the vector (the third LCC bit).This process employs convolutional encoding that converts the binaryinput symbol (0 or 1) to another symbol (0, 1, 2, or 3) and maps theencoded symbol to its corresponding QPSK signal constellation point. Theoutput of the trellis encoder is therefore a vector of 51 elements whereeach element is a signal within a set of QPSK constellation signals. The51-element vector is divided into three 17-element vectors. An LMP isadded to each of the vectors to form three (18×1) vectors d_(rev),d′_(rev), and d″_(rev), with the LMP as the first element of thesevectors. Each resulting vector is pre-multiplied by a (18×18) reversesmearing matrix C_(rev-smear) to yield another three (18×1) vectors b,b′, and b″. Each vector is post-multiplied by its corresponding (1×4)reverse spreading vector (g^(H) _(rev), g′^(H) _(rev), or g″^(H) _(rev))to yield three (18×4 matrices S_(rev), S′_(rev), and S″_(rev). Elementsof matrix S_(rev) are mapped to tones within traffic partition Aaccording to the mapping discussed in FIG. 22 and are sent to the lowerphysical layer. Similarly, elements of matrices S′_(rev), and S″_(rev)are mapped to tones within traffic partition B and C, respectively, andare sent to the lower physical. The lower physical layer places thebaseband signals in their corresponding DFT frequency bins where thedata is converted into the time domain and sent for transmission overthe air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next reverse channel transmission burst.The various steps in the transformation of binary data are shown in FIG.62.

The encryption function is identical to that for the forward channeldescribed herein.

The trellis encoding schemes for all three capacity modes are identicalto those for the forward channel described herein.

This section specifies the smearing matrix C_(rev-smear). The input tothe smearing block is the (18×1) vector D_(rev). The output of thesmearing operation (vector b) can then be described by the matrixmultiplication of d_(rev) and the (18×18) smearing matrix C_(rev-smear).That is

b=C _(rev=smear) d _(rev)

C_(fwd-smear) is the constant valued matrix shown below:

1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 α ^(βδ0) 0 0 0 0 0 0 0 0 0 0 0 0 0 00 0 α 0 ^(βδ1) 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 α 0 0 ^(βδ2) 0 0 0 0 0 0 00 0 0 0 0 0 0 α 0 0 0 ^(βδ3) 0 0 0 0 0 0 0 0 0 0 0 0 0 α 0 0 0 0 ^(βδ4)0 0 0 0 0 0 0 0 0 0 0 0 α 0 0 0 0 0 ^(βδ5) 0 0 0 0 0 0 0 0 0 0 0 α 0 0 00 0 0 ^(βδ6) 0 0 0 0 0 0 0 0 0 0 α 0 0 0 0 0 0 0 ^(βδ7) 0 0 0 0 0 0 0 00 α 0 0 0 0 0 0 0 0 ^(βδ8) 0 0 0 0 0 0 0 0 α 0 0 0 0 0 0 0 0 0 ^(βδ9) 00 0 0 0 0 0 α 0 0 0 0 0 0 0 0 0 0 ^(βδ10) 0 0 0 0 0 0 α 0 0 0 0 0 0 0 00 0 0 ^(βδ11) 0 0 0 0 0 α 0 0 0 0 0 0 0 0 0 0 0 0 ^(βδ12) 0 0 0 0 α 0 00 0 0 0 0 0 0 0 0 0 0 ^(βδ13) 0 0 0 α 0 0 0 0 0 0 0 0 0 0 0 0 0 0^(βδ14) 0 0 α 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 ^(βδ15) 0 α 0 0 0 0 0 0 0 00 0 0 0 0 0 0 0 ^(βδ16)

where,

a=(r _(LMP)/(1+r _(LMP)))^(1/2)

b=(1/(1+r _(LMP)))^(1/2)

r_(LMP) is the ratio of pilot to data power that is a physical layerprovisionable parameter whose value is nominally set to one.

The d_(i)s are elements of the cluster scrambling vector d_(smear) thatis unique to the Remote Unit. d_(smear) is a 17-element vector that isused to ensure that the smeared data from one user received in aparticular traffic partition at the Base is uncorrelated with otherusers within the same traffic partition in the local spatial cell andadjacent spatial cells, d_(smear) is given by: ${\quad \begin{matrix}d_{0} \\d_{1} \\\vdots \\d_{15} \\d_{16}\end{matrix}\quad }$

or ${\quad \begin{matrix}{{ei}\quad {\varphi (0)}} \\{{ei}\quad {\varphi (1)}} \\\vdots \\{{ei}\quad {\varphi (15)}} \\{{ei}\quad {\varphi (16)}}\end{matrix}\quad }$

The ith element of d_(smear) has the form e^(if) _(smear) ^((i))where_(fsmear)(i) is a real number between 0 and 2p generated by apseudo-random number generator creating unique sequences for each RemoteUnit. The details of the pseudo-random number generator areimplementation dependent and need not be known at the Base.

Spectral Spreading

This section defines the (1×4) reverse spectral spreading vector g^(H)_(rev) shown in FIG. 57. The input to the spectral spreading block isthe (18×1) vector b. The output of the spectral and spatial spreadingoperation, (18×4) matrix S_(rev), is the matrix multiplication of b andthe (1×4) Spectral spreading vector g^(H) _(rev):

S _(rev) =bg ^(H) _(rev)

where,

g ^(H) _(rev) =[g 0 g 1 g 2 . . . g 30 g 31]

The elements of vector g^(H) _(rev) are transmit spreading weightscalculated throughout the transmission. The algorithm for the derivationof these weights is implementation dependent. However, to clarify theprocedure a specific algorithm for the derivation of these weights isdescribed below.

The Remote Unit derives its new transmit weights based on the mostrecent data received on the forward channel. The transmit weights are ascaled version of the received weights using four receive frequenciesfor a single antenna.

The receive weight vector w^(H) _(fwd) has four elements (w₀−w₃) thatare mapped to spectral components as shown in FIG. 63.

For the Remote Unit traffic establishment procedure, the transmitweights (g₀−g₃) are calculated according to the following equation:

g ^(H) _(rev)(p)=a _(rev)(n)p _(rev) w ^(H) _(fwd)

where a_(fwd)(n) is the Base gain ramp-up factor for the nth packet andwhere p_(rev) is the Remote Unit power management factor defined by theequation below:

p _(rev) =l _(p) k _(fwd)+(1−l _(p))k _(rev)(p _(loss)(n,p)/∥(w_(fwd)(p))∥

where,

l_(p) is the exponential decay or “forget factor” nominally set to 0.97

p_(loss) is the reciprocal of the Base-Remote Unit channel gain measuredusing the Remote Unit Synchronization Pilot (RSP) tones

k_(rev) is the target Base receive power (nominally −103 dBm)

n is the burst index

p is the link index

For the Remote Unit traffic establishment procedure, the receive weightsare adaptively calculated using the following equation:

w _(fwd) =R ⁻¹ _(xx) r _(xd)

where

w_(fwd) is the (4×1) receive weight vector

r_(xd) is an estimate of the (4×1) cross-correlation vector of thereceived (4×1) vector x and the LMP (or the desired data) d

R⁻¹ _(xx) is an estimate of the (4×4) inverted auto-correlation matrixof the received vector x

For the Remote Unit steady-state procedure, receive weights areadaptively calculated using the following equation:

w _(fwd) =R ⁻¹ _(xx) r _(xy)

where

w_(fwd) is the (4×1) weight vector

r_(xy) is an estimate of the (4×1) cross-correlation vector of thereceived (4×1) vector x and the despread data y

R⁻¹ _(xx) is an estimate of the (4×4) inverted auto-correlation matrixof the received vector x

The receive weights (w₀−w₃) are mapped to spectral components accordingto the mapping shown in FIG. 63. The transmit weights (g₀−g₃) are ascaled version of the receive weights. The scaling is made according tothe following equation:

g ^(H) _(rev)(p)=p _(rev) w ^(H) _(fwd)

where p_(rev) is the Remote Unit power management factor definedearlier.

Correlation estimates are computed over four forward-channel burst. Thenew despreading weights are applied to four forward channel bursts withno delay. The spreading weights are applied to eight reverse channelbursts after an 8-burst delay. Correlation estimates are made using anexponentially average block summation. The exponential decay constant isprovisional with a nominal value of 0.7.

Reverse Control Channel Transmission Format

The block diagram for the physical layer of the solicited andunsolicited Common Access Channel (CAC) channel transmissions is shownin FIG. 64.

A CAC message is a 56-bit binary sequence composed of a trainingsequence, information bits, and CRC parity bits. The vector formationblock converts the binary sequence into a (56×1) vector. Each element ofthe resulting vector is mapped into its corresponding signal in the BPSKsignal constellation to form another (56×1) vector s. The mapping forthe BPSK signal is shown in FIG. 65.

The resulting vector is passed through two parallel paths. In the firstpath, the vector s is sent directly for spectral spreading that involvespost-multiplying it by the (1×2) spectral spreading vector g_(cac) ^(H):

g ^(H) _(cac)=[1 1]

The resulting (56×2) matrix is D_(cac) given by:${\quad \begin{matrix}{s(0)} & {s(0)} \\{s(1)} & {s(1)} \\{s(54)} & {s(54)} \\{s(55)} & {s(55)}\end{matrix}\quad }$

where s(k) is the kth element of vector s. Matrix D_(cac) is then sentto the demultiplexer where it is partitioned (by group of eight rows)into seven (8×2) submatrices D₀ to D₆.

In the second path, the vector s is code-gated. The code-gatingoperation is described by the element-wise multiplication of the (56×1)vector s with a (56×1) code-gating vector Y_(cac). The resulting (56×1)vector is s′:

s′=s·i _(cac)

The vector i_(cac) is described below.

The resulting (56×1) vector s′ is sent for spectral spreading thatinvolves post-multiplying it by the (1×2) spectral spreading vectorg_(cac) ^(H). The resulting (56×2) matrix is D′_(cac):${\quad \begin{matrix}{s^{\prime}(0)} & {s^{\prime}(0)} \\{s^{\prime}(1)} & {s^{\prime}(1)} \\\cdots & \cdots \\{s^{\prime}(54)} & {s^{\prime}(54)} \\{s^{\prime}(55)} & {s^{\prime}(55)}\end{matrix}\quad }$

where s′(k) is the kth element of vector s′. Matrix D′_(cac) is thensent to the demultiplexer where it is partitioned (by groups of eightrows) into seven (8×2) submatrices D′₀ to D′₆.

The transmission of one 56 bit CAC message requires 14 reverse channelbursts. In each of these burst, one of the 14 (8×2) matrices is mappedonto tones and sent to the lower physical layer for transmission overthe air. The interleaving and tone mapping functions are describedbelow.

The code-gating vector i_(cac) is:

i _(cac) =b _(cac) ·h _(cac)

and

b _(cac)(I)=e ^(j2pik)offset^(/56)

where b_(cac)(i) is the ith element of the (56×1) vector b_(cac). Thek_(offset) is the BSOC for the receiving Base, that ranges between 0 and31. Every Remote Unit is assigned a pair of code keys: the solicited CACcode key and the unsolicited CAC code key. The code keys are integernumbers between 0 and 63.

The 56 elements of the vector h_(cac) are the first 56 elements of thekth column of the (64×64) Hadamard matrix (H₆₄), where k is the value ofthe solicited or the unsolicited code key for the transmitting RemoteUnit depending on the type of CAC transmission. For instance, if, for agiven Remote Unit, the solicited code key is the number 13, and theunsolicited code key is the number 15:

In SCAC transmissions, elements of the vector h_(cac) are the first 56elements of the 13th column of the (64×64) Hadamard matrix

In UCAC transmissions, elements of the vector h_(cac) are the first 56elements of the 15th column of the (64×64) Hadamard matrix

There are 14 burst in every CAC transmission (burst 0 to burst 13). Theinterleaver outputs one of the 14 possible (8×2) matrices (D₀ to D₆) or(D′₀ to D′₆), in each burst. FIG. 66 shows the order of the transmissionused by the interleaver. There are two CACs in each sub-band pair. TheRemote Unit will use one of these channels depending on the CAC IDparameter received from its MAC layer. If the CAC ID is 0, the CAC_(i,0)is selected; if the CAC ID is 1, the CAC_(i,1) is selected. FIG. 67shows the mapping of the (8×2) matrix at the output of the interleaverinto tones.

Lower Physical Layer Format

The transmitter for the lower physical layer of this embodiment isfunctionally described by the block diagram in FIG. 68. The lowerphysical layer functionality is identical in forward and reversechannels.

In the forward channel, for traffic channel transmissions, the processshown in FIG. 68 is performed in parallel eight times for eightdifferent antenna elements. Furthermore, the Base may combine dataintended for various users into the same DFT bin to reduce processingrequirements. It is possible to further reduce the processing bysimultaneously transmitting traffic and control information (at the Baseor the Remote Unit) as they are carried on non-overlapping frequencytones. These techniques, however, are implementation dependent and donot change the functional characteristics of the DFT operation. As shownin FIG. 68, complex baseband signals enter the tone mapping block, wherethey are assigned into tones according to a unique mapping to either atraffic or a control channel.

The tone-mapped complex signals are complex signals are demultiplexedinto lower sub-band and upper sub-band tones, and are placed into theircorresponding DFT bins. The remaining DFT bins are filled with zeros andthe inverse DFT operation is performed, thereby transforming the datainto the time domain. The discrete time-domain samples are thenconverted into an analog signal, converted to the appropriate RFfrequency, and transmitted over the antenna.

As there are four sub-band pairs, there are four pairs of DFT blocks,where each DFT block spans one MHz of usable bandwidth. The spacingbetween the adjacent bins in one DFT block is 3.125 kHz. Each DFT blockhas 512 bins of which only 320 bins are used. Tone mapping intocorresponding DFT bins in each DFT block are shown in FIG. 69. FIG. 70depicts tone mapping pictorially. As shown, the frequency span of oneDFT block is 1.6 MHz where only 1 MHz is used for data transmission. Therelationship between tones and the actual frequency for each bin isexplained herein.

The inverse DFT operation is carried out to convert the baseband signalsinto time domain. The mathematical representation of the operation is:

x(n)=S X(k)^(ej2pnk/512)

where X(k) is the complex baseband signals in the frequency domain (thecontents of the kth bin of a DFT block), and x(n) is the nth real-valuedcomponent of the time-domain sample. The inverse DFT operation may becarried out using Inverse Fast Fourier Transform (IFFT) techniques.

The baseband transmit signals obtained after the IDFT operation must bereal. The real-valued time-domain sample outputs are then converted tothe proper RF frequency and the appropriate analog waveform fortransmission.

Airlink Physical Layer Power Output Characteristics

The power output characteristics of Base transmissions on the forwardchannel are different from that of the Remote Unit transmissions on thereverse channel.

The forward channel transmission from a Base to a given Remote Unit ismaintained at a fixed power level during the duration of a connection.The power level is determined by the Base radio management entity (RME)prior to the start of the connection using a power management algorithm.

A forward RF channel transmission is initiated by a 180 ms ramp-upperiod (240 forward channel bursts) during the traffic establishmentperiod. The ramp-up starts after a connection is established between theBase and a given Remote Unit. The data transmitted during this periodare known link maintenance pilots. The maximum (steady state) power isreached after 240 channel bursts (180 msec) and maintained throughoutthe connection.

The following equation shows the forward channel ramp-up schedulerelative to the steady state power,

a _(fwd)(n)=(1−e ^(−5(8[n/8]))/(1−e ⁻⁵))2

for n<240

a_(fwd)(n)=1

otherwise

where n is the forward channel burst number relative to the start of thetransmission.

The reverse channel transmissions from a Remote Unit to its Base isadaptively varied to ensure that the received power from all RUs attheir Base is maintained at a relatively constant level. The Remote Unitpower management algorithm is implementation dependent. One example ofthe algorithm is discussed in the Section on the Reverse Channel Format

A reverse RF channel transmission is initiated by a 180 ms ramp-upperiod (240 reverse channel bursts) during the traffic establishmentperiod. The ramp-up starts after a connection is established between theRemote Unit and its Base. The data transmitted during this period areknown LMPs. The maximum (steady state) power is reached after 240reverse channel bursts (180 msec).

The following equation shows the reverse channel ramp-up schedulerelative to the steady state power,

a _(rev)(n)=(1−e ^(−5(8[n/8]))/(1−e ⁻⁵))2

for n<240

a_(rev)(n)=1

otherwise

where n is the reverse channel burst number relative to the start of thetransmission.

A “Proof-of-Concept” Embodiment

The signal processing procedure described generally above can beimplemented in a “proof-of-concept” embodiment by circuitry within thehigh-bandwidth base station 110 and the radio access stations 187, 192.In addition, the dynamic bandwidth allocation method of the presentinvention is implemented in a “proof-of-concept” embodiment within thecircuitry of the communications network 100 depicted below.

FIG. 71 is a schematic block diagram showing the main structuralelements of one implementation of the high bandwidth efficiency,bandwidth-on-demand communications network 100. Specifically, thecommunications network 100 is shown to include a plurality of full-rate,high-bandwidth, radio access stations 192 as well as a low-rate,high-bandwidth radio access station 187. Typically, a full-rate,high-bandwidth radio access station 192 is able to provide forcommunication between a base station 110 and a large number ofsubscribers 130, while the low-rate, high-bandwidth radio access station187 is able to provide for communication with the base station 110 onlyone or a few subscribers 130 at a time.

The subscribers 130 communicate with the full-rate or low-rate,high-bandwidth radio access stations 192, 187 via a cable or othercommunication link. The high-bandwidth radio access stations 192, 187,in turn, communicate bidirectionally with the base station 110 viawireless communications channels to form an air-link. The structure andoperation of the base station 110 as well as the structure and operationof the full and low-rate, high-bandwidth radio access stations 192, 187,will be discussed in greater detail below with reference to FIG. 72 and73.

The base station 110 together with the full and low-rate, high-bandwidthradio access stations 192, 187 together comprise a subsystem 150. Thesubsystem 150 communicates bidirectionally with a telecommunicationsnetwork 160 via a land line 170 that may, for example, comprise a coppercable or a fiber-optic connection. Alternatively, the link 170 maycomprise a microwave link. The telecommunications network 160 mayinclude for example, the public switched telephone network, a mobiletelephone switching office (MTSO), a private data network, a modem bankor a private branch exchange, as is well known in the art.

FIG. 74 is a simplified schematic block diagram that shows the mainfunctional and structural elements of the bandwidth-on-demandcommunications network 100 in greater detail. The communications network100 is shown in FIG. 74 to connect a plurality of the subscriber units(e.g., the computer 131, the telephone 132, a plurality of telephones140 in communication with a public switching network, or a plurality ofcomputer terminals 145 within a local area network) to public or privatedata or telephone networks 150-156. Public data network 150, the privatedata network 152, the private telephone network 154, and the publictelephone network 156 communicate with an asynchronoustelecommunications multiplexer (ATM) 162 over lines 163, 164, 165, and166, respectively, via a plurality of network interfaces, designatedgenerally by a block 160. The asynchronous telecommunicationsmultiplexer 162 acts as a multiplexing switch and connects with thehigh-bandwidth base station 192 via the communications link 170, thatadvantageously comprises a fiberoptic link, a copper wire, or amicrowave transmission link. The high-bandwidth base station 110provides radio frequency output signals to the receiving stations viathe antenna 120.

A bandwidth demand controller 175 communicates with the high-bandwidthbase station 110, asynchronous telecommunications manager switch 162,and the network interfaces 160 via lines 176, 177 and 178, respectively.The bandwidth demand controller 175 also communicates with anintelligent service node 180 via a line 179. The intelligent servicenode 180 communicates with the ATM switch 162 via a line 182. Theabove-described elements of the bandwidth on demand communicationsnetwork 100 comprise the telecommunications network side 183 of thebandwidth-on-demand communication system 100. The telecommunicationsnetwork side 183 communicates with the low rate high-bandwidth radioaccess station 187 via an antenna 185 or the fill rate high band widthradio access station 192 via an antenna 190. The radio access station187 connects to a plurality of the subscriber units including thetelephone 132 and the computer 131. The radio access station 192 isconfigured to communicate with multiple subscribers 140 via a publicswitching network 195 that connects to the radio access station 192 viaa communications link 194. The full rate radio access station 192further connects to the computer terminals 145 via a local network 197that connects to the radio access station 192 via communication link196. Each of the subscriber units 131, 132, 140 and 145, together withthe elements 185-197 of the communications system 100, comprise asubscriber network side 199 of the bandwidth-on-demand communicationssystem 100.

The high-bandwidth base station 110 in association with the bandwidthdemand controller 175, and the high-bandwidth radio access stations 187,192 that communicate with the high-bandwidth base station 110 via an airlink, are the heart of the bandwidth on demand communication system 100.Although only a single high-bandwidth base station 100 is depicted inFIG. 74, it will be understood that a plurality of high-bandwidth basestations are advantageously included within the high-bandwidthcommunication system 100. Each of the high-bandwidth base stations 110is capable of supporting from one to hundreds of simultaneousbi-directional users. Each user may request in advance, or optionally,during the course of communications, an amount of bandwidth from 8kilobits to 1.544 megabits per second. Furthermore, each high-bandwidthbase station 110 may have one or multiple transmitting and receivingantennas 120. The high-bandwidth base stations 110 are high-bandwidthradio transceivers, that, with their associated antennas 120, may belocated on towers, on top of buildings, inside buildings, or in otherconvenient locations.

A bandwidth controller 175 is associated with the high-bandwidth basestations 110. The bandwidth demand controller 175 provides intelligenceto monitor information transmitted to the base stations 110 from theradio access stations 187, 192. Specifically, the informationtransmitted from the high-bandwidth radio access stations 187, 192 areconverted to intelligence within the bandwidth demand controller 175 inorder to instruct the base stations 110 how much bandwidth to provide agiven radio access station 187, 192. Although shown in FIG. 74 as aseparate element from the high-bandwidth base station 110, the bandwidthdemand controller 175 may be integral to a base station 110, may beattached locally to a base station 110, or may be remote and connectedto a base station 110 via the communication link 176. The bandwidthdemand controller 175 further acts as a central bandwidth controllerthat insures that the bandwidth appropriated at each communication linkthroughout the communications network 183 is consistent with thebandwidth assigned to a particular channel on the high-bandwidth basestation 110. Thus, the bandwidth demand controller 175 controlsbandwidth allocated within the asynchronous telecommunicationsmultiplexer switch 162, and the network interfaces 160. Additionally,the bandwidth demand controller 175 communicates bandwidth informationto the intelligent service node 180 that is used to manage the deliveryof the user data to the appropriate network 150-156. The intelligentservice node 180 can control the ATM switch 162 to manage bandwidthchanges and the network interfaces 160.

The high-bandwidth radio access stations 187, 192, as shown in FIG. 74,are exemplary of a plurality of high-bandwidth radio access stationsthat are included within the bandwidth on demand communication system100. One or more of the high-bandwidth radio access stations 187, 192are capable of communicating with one or more high-bandwidth basestations 110 utilizing the air interface. In addition, each of the radioaccess stations 187, 192 is capable of supporting one or more interfacessuch as the connections between the telephone 132 and the standardcomputer 131, as well as the telephone network interface (PBX) 195 andthe computer network comprising the terminals 145 and the LAN 197. Thehigh-bandwidth radio access stations 187, 192 have the capability tointerpret bandwidth needs of the devices connected to the radio accessstations 187, 192, and communicate these bandwidth needs via the airinterface and the base station 110, to the bandwidth demand controller175. Advantageously, the bandwidth demand controller 175 can furthercommunicate these bandwidth demands to the ATM switch 162 or theintelligent service node 180, and the network interfaces 160.

In operation, one of the connected subscriber units (e.g., the computer131, the telephone 132, the PBX 195, or the LAN 197) requests bandwidthsvia a connection to one of the high-bandwidth radio access stations 187,192. The radio access station 187, 192 transmits a request for accessand bandwidth to the high-bandwidth base station 110 via the antenna185, 190, the air interface, and the antenna 120. The request for accessis made via a communications control channel available to allsubscribers within the area of use. If two subscribers simultaneouslyrequest connection, then a random accessing protocol is employed todetermine which unit is first granted control of the communicationscontrol channel.

The high-bandwidth base station 110 communicates all bandwidth requeststo the bandwidth demand controller 175. The bandwidth demand controllerperforms an allocation of the requested bandwidth and advantageouslyarranges system resources within the telecommunications network side 183(including the intelligent service node 180, the ATM switch 162, and thenetwork interfaces 160). Once the bandwidth demand controller 175determines the amount of bandwidth available for allocation, andcompares this with the requested bandwidths, the bandwidth demandcontroller 175 either immediately allocates the requested bandwidth, orbegins a negotiation process using the available amount of bandwidth.This bandwidth negotiation occurs between the bandwidth demandcontroller and the radio access station 187, 192 through the basestation 110 and the air interface.

Thus, the radio access station 187, 192 either receives anacknowledgment that the bandwidth requested is available, andsubsequently begins transmitting data, or the radio access station 187,192 receives an offer of less bandwidth from the bandwidth demandcontroller 175. If an offer of less bandwidth is transmitted to thehigh-bandwidth radio access station 187, 192, the radio access station187, 192 determines whether the connected device or network caneffectively operate with the offered bandwidth. If the connected deviceor network can effectively operate with the offered bandwidth, the radioaccess station 187, 192 begins transmitting data at the offeredbandwidth. However, if the radio access station 187, 192 determines thatthe offered bandwidth is not adequate for operation of the connecteddevice or network, the radio access station 187, 192 notifies theconnected device or network that access is not available, and furthernotifies the bandwidth demand controller 175 (via the base station 110and the air interface) that the offered bandwidth will not be used bythe radio access station 187, 192.

If a suitable bandwidth is available, the bandwidth controller allocatesthis bandwidth to establish a communications channel with the requestingsubscriber. Thus, for example, the telephone subscriber unit 132 mayindicate that a data rate of 8 Kb per second is required (thatcorresponds to a particular bandwidth) while the computer subscriberunit 131 may indicate that a total transmission rate of 128 Kb persecond (corresponding to another given bandwidth) in order to establisheffective communications with the high-bandwidth base station 110. Ifthe communications network 100 is unable to provide the requested amountof bandwidth, a negotiations process commences wherein thehigh-bandwidth base station 110 transmits an alternative bandwidth, thatis less than the requested bandwidth, to the requesting subscriber unitvia the radio access station 187. The requested subscriber unit thenindicates to the high-bandwidth base station 110 whether or not theallocated bandwidth is suitable for the communications needs of thesubscriber unit.

As will be described in greater detail below, the bandwidth demandcontroller 175 allocates bandwidth by assigning one or more frequencytone set and one or more spreading code to the subscriber unit inaccordance with a pre-defined bandwidth allocation procedure. Each toneset and spreading code increases bandwidth by an additional factor. Inone advantageous embodiment, bandwidth can be allocated in amounts assmall as 8 Kbits/sec to as large as 1.544 Mbits/sec to define thecommunications channel.

Once a communications channel is established for the requestingsubscriber 130, data representing either human voice communications orcomputer-to-computer communications in digital form, is transmittedbetween the high-bandwidth base station 110 and the high-bandwidth radioaccess station 187, 192. As will be described in greater detail below,the digitally encoded signal contains forward error correction togetherwith signal spreading and other modulation techniques.

All data received by the high-bandwidth base station 110 from allcommunicating radio access stations 187, 192 are multiplexed into anasynchronous telecommunications multiplexed data stream and transmitted,via the communication link 170, to the ATM switch 162. At the ATM switch162, the data stream is switched (i.e., demultiplexed) with the optionalassistance of the intelligent service node 180 to the appropriatenetwork interfaces 160, and from there onto the appropriate network150-156.

As will be discussed in further detail below, the bandwidth demandcontroller 175 also controls bandwidth allocation for the networkinterfaces 160 and the ATM switch 162. In this manner, the bandwidthallocated throughout an entire communications link (i.e., from asubscriber to a data or telephone network) can be flexibly assignedaccording to the needs of each subscriber unit. Furthermore, thepreferred embodiment assures that the bandwidth through the airinterface and the bandwidth through the land line connections areappropriately matched.

When a device or network connected to a high-bandwidth radio accessstation 187, 192 no longer requires bandwidth, the radio access station187, 192 ceases transmission to the base station 110, and notifies thebandwidth demand controller that the bandwidth is now released forreallocation.

The “Proof-of-Concept Embodiment”—Remote Terminal Hardware

FIG. 72 is a functional block diagram that shows the main functionalelements of the full-rate, high-bandwidth radio access station 192. Itshould be understood, for purposes of the present description, that thefull-rate, high-bandwidth radio access station 192 described herein issubstantially identical in structure and operation to the low-rate,high-bandwidth radio access station 187, with the exception that thelow-rate, high-bandwidth radio access station 187 provides communicationaccess for only a single subscriber 130. As shown in FIG. 72, thefull-rate, high-bandwidth radio access station 192 comprises a transmitreceive switch 300 that connects bidirectionally with the antenna 120.The structure and operation of the antenna 120 will be described ingreater detail below with reference to FIGS. 6 and 7. The transmit andreceive switch 300 connects to a down converter 305 when in the receivemode and an up converter 307 while in a transmit mode. Thetransmit/receive switch 300 further receives synchronization and packettiming data from a synchronization circuit 312.

The down converter 305 receives the radio signals from the antenna 120via the switch 300. In addition, the down converter 305 receives a localoscillator reference as well as an analog-to-digital converter clockfrom the synchronization circuit 312. The down converter 305communicates with a demodulator 310 that, in turn, provides a feedbackof automatic gain control level to the down converter 305. Thedemodulator 310 communicates bidirectionally with the synchronizationcircuit 312 and also provides an output to a code-nulling circuit 315.The code-nulling circuit 315 provides a frequency error signal to thesynchronization circuit 312, and also communicates with amultidimensional trellis decoder 320. The multidimensional trellisdecoder 320 connects to a digital data interface 325. The digital datainterface 325 communicates bidirectionally with a remote control circuit330. The remote control circuit 330 receives inputs from the demodulatorcircuit 310, the code-nulling circuit 315, and the multidimensionaltrellis decoder 320. The control circuit 330 further transmits statussignals and receives command signals from the base station 110 (see FIG.74). Finally, the remote control circuit 330 outputs axis parameters toa multidimensional trellis encoder 335, that also communicates with thedigital interface 325. The multidimensional trellis encoder 335communicates with a SCMA coding circuit 340. The SCMA coding circuit 340further receives an input from the code-nulling circuit 315. The SCMAcoding circuit 340 outputs signals to a modulator circuit 345 that alsoreceives an input from the synchronization circuit 312. Finally, themodulation circuit 345 together with the synchronization circuit 312provide inputs to the up converter 307. The up converter 307 outputs thedata signal to the transmit receive switch 300 while the transmitreceive switch is in the transmit mode. This signal is output over anerror interface to the multiple subscribers 130 via the antenna 120.

In operation, once it is the proper time to receive a data packet, thetransmit/receive switch 300 switches the antenna 190 into the downconverter 305. The down converter 305 takes the signal at thetransmission frequency (see, e.g., about 2 gigaHertz), and translatesthis to the proper frequency for digitization. The DMT-SC demodulatorthen performs a fast Fourier transform (FFT) and presents the individualfrequency bins to the code-nulling network 315. As discussed brieflyabove, the code-nulling network 315 applies code-nulling weights to thedespreading codes in order to cancel interference due to transmissionshaving non-orthogonal spreading codes. The code-nulling network 315 alsodespreads the demodulated signal provided by the DMT-SC demodulator 310and produces output demodulated symbols.

The demodulated symbols are provided as an input to themulti-dimensional trellis decoder 320 in order to decode the symbols inaccordance with pragmatic Viterbi decoding methods. Receive bits areprovided at the output of the multidimensional trellis decoder 320. Thereceive bits pass through a digital data interface 325 that, in oneembodiment, serves as a data interface for a T1 link.

On the transmit side, data to be transmitted enters the digital datainterface 325 via the T1 link and enters the multidimensional trellisencoder 335 for trellis encoding. It will be understood, of course, thatother kinds of error encoding and symbol encoding such as Reed-Solomonerror coding, and QAM or BPSK symbol encoding are performed within theencoder 335. The encoded symbols enter the spreading circuit 340 whereinthe spreading code together with the appropriate code weights areapplied to the input symbols. The spread symbols are DMT-SC modulated asrepresented within the block 345 and the resulting signal is translatedto the high frequency band via the up-converter 307. Thetransmit/receive switch 300 is then switched to connect the up converter307 to the antenna 190 so that the modulated and encoded data signal istransmitted via the antenna 190.

Immediately after one of the radio access terminals 187, 192 has beeninstalled and is just coming on-line for the first time, the radioaccess station 187, 192 does not have information regarding the locationof the assigned base station 110. Furthermore, the remote access station187, 192 does not have information concerning the interference resultingfrom other transmitters and reflectors within the environment of theremote station 187, 192. Thus, each remote, upon initialization, must“learn” the location of the base station as well as the location ofdifferent interferes and reflectors within the immediate environment ofthe remote. Because the remote installer points the remote antenna arrayin the direction of the nearest base station 110, the strongest signalreceived by the remote is generally from around the 0° direction. Theremote subsequently fine tunes, or adaptively adjusts the beam formingso as to obtain the maximum SINR for the signal received from thenearest base station 110.

When the radio access station 192 transmits to the base 110, the basestation 110 expects to receive each of the signals transmitted from theremotes at the same power level. Thus, a gain control level is reportedto the remote control 330 within the radio access station 192 from theDMT-SC demodulator 310. This automatic gain control level (AGCL) is alsotransmitted from the DMT-SC modulator 310 to the up-converter 307 sothat the gain of the power amplifier (not shown within the up-converter307) can be adjusted. In this manner, the base stations 110 can assurethat the signal transmitted from the remote access terminals 187, 192arrive at the base station 110 at the proper level.

The radio access stations 187, 192 also have to perform synchronization.That is, although the remote access terminals 187, 192 are preprogrammedto operate within a TDD system, the specific information concerning thedistinction between the transmit and receive packages as well as theexact timing of the packet transfer still must be determined by theradio access stations 187, 192 when a radio access station first comeson line. Subsequently, the remote terminals 187, 192 must acquirefrequency synchronization for the DMT-SC signals so that the remotes areoperating at the same frequency and phase as the base station 110. Forthis reason, the DMT-SC demodulator 310 generates a packet referencethat is utilized by the synchronization circuitry 312 to establish thebasic transmit/receive timing (i.e., the packet timing for the T/Rswitch 300). In addition, the packet timing is provided as a receivegate to the demodulator 310 and as a transmit gate to the modulator 345so that the remote access station 187, 192 transmits and receives at theappropriate intervals.

Within the code-nulling network 315, measurements are taken on thewaveform to determine the frequency error. The measured frequency erroris provided to the synchronization circuitry 312 so that the radioaccess station 187, 192 can come into frequency and phase lock with thebase station 110. This synchronization information is transmitted fromthe synchronization circuitry 312 to the up-converter 307 and thedown-converter 305 as a local oscillator reference and also as adigital-to-analog converter clock (or conversely, an analog-to-digitalconverter clock).

The code-nulling network 315 also estimates the characteristics of themultitask channel (i.e., the frequency response of the multipathchannel). The channel estimates are provided to the spreading circuitry340 so that the preemphasis function can be performed to adaptivelyequalize the multipath channel. Furthermore, the code-nulling network315 provides an estimate of the received power and the SINR to theremote control circuitry 330. In addition, an estimate of the bit errorrate (BER) is provided from the multidimensional trellis decoder 320 tothe remote control circuitry 330. These parameters are used by theremote control circuitry 330 to control the flow of data via the digitaldata interface 325 with the subscriber (e.g., a PBX or a LAN).Furthermore, status signals based upon these input parameters to theremote control circuitry 330 are also transferred to the subscribers.The status signals indicate to the subscribers whether or not the radioaccess terminal is operating properly.

When the radio access terminals 187, 192 first dials onto the network100 (i.e., the remote is trying to establish connection with the base)the base provides the remote 187, 192 with a set of access parametersthat includes, for example, the appropriate starting codes to use, whichtone sets to receive and transmit on, etc., so that a communicationchannel is set up between the base station 110 and the remote station187, 192.

FIGS. 21A and 21B depicts the digital architecture within the remoteaccess terminals 187, 192. The remote digital architecture includes aninterface card 2100 that communicates bidirectionally with a layerprocessing accelerator (LPA) card 2110 as well as a transmitting LPAcard 2120.

The interface card 2100 (shown in greater detail in FIG. 76 below)includes an ETHERNET interface card, a global positioning system (GPS)interface and other control interfaces. The ETHERNET interfacecommunicates bidirectionally with a monitoring computer such as an AppleMacIntosh, while the GPS interface derives timing data forsynchronization purposes from the base station transmission, while thecontrol interface outputs printer control bits for controlling thetuner. The interface card 2100 further includes three digital signalprocessing chips that, advantageously comprise PMS320C40 digital signalprocessing chips (“C40s”) available from Texas Instruments. In addition,a Viterbi decoder as well as a T1 and an integrated services digitalnetwork (ISDN) interface are included on the interface card 2100 toprovide an interface between the T1 communication link as well as theISDN communication link with the subscribers.

As shown in FIG. 75A, the interface card 2100 further includes anadditional PMS320C40 digital signal processing chip as well as anadditional Viterbi, T1, ISDN interface that are crossed out. This is toindicate that these chips, although physically present on the interfacecard 2100, are not used within the remote digital subsystem although thesame interface card is typically used in the base station 110. This isdone because it is less expensive to manufacture a single interface cardfor both the base station 110 and the remotes 187, 192 rather thanproviding a specific card for the remotes and bases.

Finally, the interface card 2100 includes a G-link receiver thatreceives sampled data from the receiver digital-to-analog converter anda G-link transmitter that transmits sample data to the transmitteranalog-to-digital converter.

The sample data received from the digital-to-analog converter passesthrough the G-link receiver within the interface card 2100. The G-linkreceiver provides the received waveform data to a receiving LPA card2110 (FIG. 75B). The LPA card 2110 will be described in greater detailbelow with reference to FIGS. 77A-77D. Briefly, the receiving LPA card2110 includes a pair of SHARP LH9124 (“9124s”) digital signal processingchips 2112, 2114, as well as a pair of Texas Instruments TMS320C40digital signal processing chips 2116, 2118.

The receiving LPA card 2110 demodulates the received data and providesthe demodulated data to one of the TMS320C40 DSP chips within theinterface card 2100. After further digital signal processing, the datais decoded and then transmitted to the subscriber via the T1 interface.Of course, it will be understood that if the radio access terminalcomprises one of the low-rate radio access terminals 187, then asuitable communications link other than a T1 link will connect to theinterface card 2100.

When data is to be transmitted, information supplied by the T1interface, or other communication link, enters the interface card 2100,as shown in FIG. 75A, and passes through a series of digital signalprocessing chips within the interface card 2100. The transmit dataoutput from the interface card 2100 enters a transmit LPA card 2120,that has a substantially similar architecture to the received LPA card2110. The transmit LPA card 2120 converts the transmit data intotransmit waveform data suitable to be sent to the transmitteranalog-to-digital converter via the G-link transmitter within theinterface card 2100.

FIG. 76 is a software block diagram that indicates the generalprocessing steps performed by each of the digital signal processingchips within the digital signal processing architecture of the radioaccess terminals 187, 192. Specifically, control signals are generatedby the TMS320C40 digital signal processing chips 2102, 2106, while thesymbol modulation (e.g., including trellis coded, Reed-Solomon, and QAM,BPSK, or M-ARY modulation) is performed by the digital signal processingchips 2104, 2106.

Within the receiving LPA, the 9124 digital signal processor 2112 inconjunction with the C40 digital signal processor 2116 perform theoperations relating to the fast Fourier transform. The 9124 digitalsignal processor 2114 in conjunction with the C40 digital signalprocessor chip 2118 perform the processing steps relating to thecode-nulling and adaptive equalization aspects of the present invention.In like manner, within the transmitting LPA 2120, the C40 digital signalprocessing chip 2124, together with the 9124 digital signal processingchip 2128, perform the digital signal processing steps relating to theinverse fast Fourier transform (IFFT), while DSP chips 2122 and 2126perform the signal spreading operations used to provide modulation inaccordance with the present invention.

FIGS. 78A-78C are more detailed block diagrams showing the digitalarchitecture used to support the main digital signal processing C40chips on the interface card 2100 of the remote terminals 187, 192.Several interface support circuits are employed to precondition datareceived by the digital signal processing chip 2102. In particular, areceive/transmit control interface circuit, an ETHERNET interfacecircuit, an erasable programmable logic device (EPLD) synchronizationcircuit, and a universal asynchronous receiver/transmitter (UART) serveas an interface between the DSP chip 2102 and circuitry external to theinterface card 2100. In addition, a programmable read-only memory(PROM)/random access memory (RAM), an electrically erasable PROM, areceived signal strength indicator (RSSI) input circuit and a pluralityof light-emitting diode (LED) switch drivers all communicate with theDSP chip 2102 via a common bus.

The C40 DSP chip 2104 is also supported by interface circuitry.Specifically, an integrated services digital network (ISDN) interfaceand a T1 interface provide a connection to ISDN and T1 equipment, whilea supporting Viterbi encoder/decoder, as well as a PROM/RAM, providedigital signal processing support for the DSP chip 2104 via a commonbi-directional bus.

In addition to receiving signals from the DSP chip 2104 and orderwireFFT data, the C40 DSP chip 2106 communicates bidirectionally with aPROM/RAM and a codec via a bi-directional common bus. The codec incommunication with the DSP 2106 communicates bidirectionally with anorderwire headset.

The G-link receiver provides a clock synchronization signal to theG-link transmitter, as well as to an EPLD. In addition, the receivingG-link transmits RSSI data to the RSSI input in communication with theC40 DSP chip 2102. The EPLD that receives the synchronization signalfrom the receiving G-link circuit provides a receive address, a framesync signal, and a transmit address as control outputs.

In operation, each of the DSP chips 2102, 2104, 2106 uses the localPROM/RAM for storage and retrieval of data and for use as a look-uptable. The C40 DSP chip 2102 receives the RSSI input data to implementautomatic game control (AGC). That is, an indication of the signalintensity is provided via the RSSI input to the DSP chip 2102 so thatthe remote terminal 187, 192 can automatically adjust the receive gainso that the signal is received at the appropriate level. The ETHERNETinterface allows the remote terminal 187, 192 to transmit data out to alocal computer or operator. The receive/transmit control interfacecircuit sends control bits to the radio frequency electronics of theremote terminal 187, 192 in order to control the RF electronics. TheEPLD synchronization circuit receives an envelope detector output fromthe RF circuit within the receiver of the remote terminal 187, 192 inorder to achieve TDD synchronization. The UART circuit provides for theinput of a universal global positioning system (GPS) time clock for useby the remote terminals 187, 192. Finally, the electrically erasablePROM allows the radio access terminals 187, 192 to store informationfrom test to test as a kind of statistical record.

The operations of the other support circuitry depicted in FIGS. 78A-78Care well known to those of ordinary skill in the art and need not bedescribed in detail for a complete understanding of the presentinvention.

FIGS. 77A-77D are a more detailed block diagram of the LPA cards 2110,2120 of the remote terminals 187, 192 that shows the support circuitryused to support the operation of the SHARP LH9124 DSP chips, as well asthe TMS320C4 DSP chips from Texas Instruments. It should be understoodthat although FIGS. 77A-77D depict only the receiving LPA card 2110 thatthe architecture of the LPA card 2110 is substantially similar to thatof the transmitting LPA card 2120 so that essentially the samedescription applies to both LPA cards. Input data in quadrature form(e.g., 24 In-phase bits and 24 Quadrature bits) are provided as an inputto a double buffer 2402 via a 48-bit input bus. A first portion of thedouble buffer 2402 is controlled via input address and control bits,while a second portion of the double buffer 2402 is controlled via anaddress generator 2404. The address generator 2404 communicates with theTMS 320C40 DSP chip 2116 via a bus 2406.

The double buffer 2402 communicates with the SHARP LH9124 digital signalprocessing chip 2112 via a bi-directional bus and also supplies data asan input to a first in/first out (FIFO) buffer 2408. In one preferredembodiment, the FIFO comprises a 5K×48-bit buffer. The FIFO 2408communicates with the DSP chip 2112, as well as with a double buffer2410. Like the double buffer 2402, the double buffer 2410 advantageouslycomprises a pair of 32K×48-bit RAMs. Furthermore, the double buffer 2410is under the control of the address generator 2412 that communicateswith the buffer 2406. The double buffer 2410 communicatesbidirectionally with the DSP chip 2116 via the bus 2406.

The SHARP DSP chip 2112 further receives input from a sine/cosinelook-up table 2414. The sine/cosine look-up table 2414 receives inputfrom a rectangular-to-polar converter 2416 that in one embodimentcomprises a signal processing chip sold under Model Number PDSP16330 andavailable from GEC Plessey. Finally, the DSP chip 2112 receivessequencing data from a sequencer 2418, that also communicates with thebus 2406. The output of the digital signal processor chip 2112 isprovided as an input to a double buffer 2420, that is substantiallysimilar in structure to the double buffers 2402 and 2410. A firstportion of the double buffer 2420 is under the control of an addressgenerator 2422 that receives signals from the DSP chip 2116 via the bus2406.

The second half of the LPA card 2110 is substantially similar inarchitecture to the first half described above. Specifically, input datain quadrature form (e.g., 24 In-phase bits and 24 Quadrature bits) areprovided as an input to a second half of the double buffer 2420 from thefirst half of the buffer 2420. The second half of the double buffer 2420is controlled via an address generator 2424. The address generator 2404communicates with the TMS 320C40 DSP chip 2118 via a bus 2426.

The double buffer 2420 communicates with the SHARP LH9124 digital signalprocessing chip 2114 via a bi-directional bus and also supplies data asan input to a first in/first out (FIFO) buffer 2428. In one preferredembodiment, the FIFO 2428 comprises a 5K×48-bit buffer. The FIFO 2428communicates with the DSP chip 2112, as well as with a double buffer2430. Like the double buffer 2420, the double buffer 2430 advantageouslycomprises a pair of 32K×48-bit RAMs. Furthermore, the double buffer 2430is under the control of the address generator 2432 that communicateswith the buffer 2426. The double buffer 2430 communicatesbidirectionally with the DSP chip 2118 via the bus 2426.

The SHARP DSP chip 2114 further receives input from a sine/cosinelook-up table 2434. The sine/cosine look-up table 2434 receives inputfrom a rectangular-to-polar converter 2436 that in one embodimentcomprises a signal processing chip sold under Model Number PDSP16330available from GEC Plessey. Finally, the DSP chip 2114 receivessequencing data from a sequencer 2438, that also communicates with thebus 2426. The output of the digital signal processor chip 2114 isprovided as an input to a buffer 2440, that advantageously comprises a32K×48 RAM. The buffer 2440 is under the control of an address generator2442 that receives signals from the DSP chip 2118 via the bus 2426.

The C40 DSP chips 2116 and 2118, respectively, receive GPS timing viaUART circuits 2450, 2452. Furthermore, each of the DSP chips 2116, 2118communicates with respective RAM chips 2454, 2456, that advantageouslycomprise 128K×32 random access memories.

The DSP chips 2116, 2118 further communicate with EPROMs 2460, 2470,respectively, and RAMs 2462, 2472, respectively, via local buses 2464,2474, respectively. In one advantageous embodiment the EPROMs 2460, 2470comprise a 512K×8 memory, while the RAMs 2462, 2472 comprise a 128K×32RAM. A pair of internal communication ports provide for communicationbetween the DSP circuits 2116, 2118, while two pair of input/outputexternal communication ports connect to each of the DSP chips 2116,2118.

In operation, the DSP chips 2116, 2118 employ the respective memories2460, 2462, 2470, 2472 to perform processing associated with the fastFourier transform and code spreading or code-nulling processingoperations. Meanwhile, the double buffer 2402 collects input datasymbols in quadrature. The double buffer 2402 is provided so that whiledata is being collected from one packet, data from the previous packetcan be processed.

As can be seen from FIGS. 77A-77D, two substantially identicalprocessing engines are provided separated by the double buffer 2420. Inone advantageous embodiment each of the 9124 DSPs 2112, 2114 operate ata 40-mHz sample rate and include six multipliers so that data can bestreamed through in substantially real time.

The “Proof-of-Concept Embodiment”—Base Station Hardware

FIG. 73 is a functional block diagram showing the main functionalelements of the base station 110 shown in FIG. 74. As shown in FIG. 73,the base station 110 includes a transmit/receive switch 400 thatcommunicates bidirectionally with a plurality of the antennas 120. Whilein the receive mode, the switch 400 communicates with a down converter405, and in the transmit mode, the transmit receive switch 400communicates with an up converter 407. The down converter 405 alsoreceives inputs from a frequency reference circuit 409 and providesoutputs to a demodulator 410. The demodulator 410 feeds back anautomatic gain control level to the down converter 405 and also receivesinputs from a packet timing generator 412. The packet timing generator412 receives analog-to-digital converter clock inputs from the frequencyreference circuit 409.

The demodulator 410 provides inputs to a beam forming and code-nullingcircuit 415. The beam forming and code-nulling circuit 415 communicateswith a multidimensional trellis decoder 420 that, in turn, communicatesbidirectionally with a network/data interface circuit 425.

The network/data interface circuit 425 provides outputs to and receivesinputs from the telecommunications network 160 (see FIG. 74).Furthermore, the network/data interface circuit 425 provides an outputsignal to the packet timing generator 412 and also communicatesbidirectionally with a base control circuit 430. The base controlcircuit 430 receives inputs from the demodulator 410, the beamforming/code-nulling circuit 415, and the multidimensional trellisdecoder 420. The base control circuit 430 also communicatesbidirectionally with an operator station (not shown) within thetelecommunications network 160.

The network/data interface circuit 425 communicates with amultidimensional trellis encoder 435. The multidimensional trellisencoder 435 provides an output to a retroactive beam forming network andSCMA circuit 440. The network 440 also receives inputs from the beamforming/code-nulling circuit 415 as well as the base control circuit430. The retroactive beam forming and SCMA network 440 provides anoutput to a modulator 445 that also receives inputs from the packettiming generator 412. Finally, the modulator 445 together with thefrequency reference circuit 409 provide inputs to the up converter 407,that in turn provides an output to the transmit/receive switch 400 whilein the transmit mode. Signals provided by the up converter aretransmitted to the various high-bandwidth radio access stations 192, 187by means of the antennas 120.

The operation of a base station is substantially similar to theoperation of the radio access station 187, 192. Specifically, thetransmit/receive switch 400 switches the antenna array 120 into thedown-converter 405. The down-converter 405 takes the signal at thetransmission frequency (see, e.g., about 2 gigaHertz), and translatesthis to the proper frequency for digitization. The multi-sensor DMT-SCdemodulator 410 then performs a fast Fourier transform (FFT) andpresents the individual frequency bins to the beam forming/code-nullingnetwork 415. As discussed briefly above, the code-nulling network 415applies code-nulling and beam forming weights to the despreading codesin order to cancel interference due to transmissions havingnon-orthogonal spreading codes. The code-nulling network 415 alsodespreads the demodulated signal provided by the multi-sensor DMT-SCdemodulator 410 and produces output demodulated symbols.

The demodulated symbols are provided as an input to themulti-dimensional trellis decoder 420 in order to decode the symbols inaccordance with pragmatic Viterbi decoding methods. Receive bits areprovided at the output of the multidimensional trellis decoder 420. Thereceive bits pass through a digital data interface 425 that, in oneembodiment, serves as a data interface for a T3/SONET interface link.

On the transmit side, data to be transmitted enters the digital datainterface 425 via the T3/SONET link and enters the multidimensionaltrellis encoder 435 for trellis encoding. It will be understood, ofcourse, that other kinds of error encoding and symbol encoding such asReed-Solomon error coding, and QAM or BPSK symbol encoding are performedwithin the encoder 435. The encoded symbols enter thebeam-forming/code-spreading circuit 440 wherein the spreading codetogether with the appropriate beam forming and null-steering codeweights are applied to the input symbols. The spread symbols are DMT-SCmodulated as represented within the block 445 and the resulting signalis translated to the high frequency band via the up-converter 407. Thetransmit/receive switch 400 is then switched to connect the up converter407 to the antenna array 120 so that the modulated and encoded datasignal is transmitted via the antenna 120.

For synchronization of the base stations 100 all of the bases 110 arelocked onto GPS time. In this manner, no matter how big thecommunications network 100 becomes, all of the base stations 110 alwayshave the proper TDD synchronization. Thus, the base stations 110 alwaysstart transmitting at the same time and receiving at the same time. Atthe packet timing generator 409, the frequency reference is GPS derivedand this is used to control the T/R switch 400. This is particularlyadvantageous because the timing does not have to be derived from thewaveforms transmitted by several remotes. Since the remote terminals187, 192 derive their synchronization timing from the base stations 110,the remotes will be synchronized to GPS time.

The packet timing generator 412 receives a clock signal from the timinggenerator 409 so that the packet timing generator 412 can supplytransmit and receive gating signals to the modulator 445 and thedemodulator 410, respectively.

In an alternative embodiment, it would be possible to establish auniversal timing mechanism for all of the base stations 110 and theremote terminals 187, 192 provided from the network via the networkinterface 425. In such an embodiment, an specially defined ATMadaptation layer could be used to provide a clock to the interface 425.Management information and connection power control information couldalso be supplied over the T3 or SONET link. Such information could beprovided to the base controller 430 that will send the proper signalsout to the remotes 187, 192 through the wireless signaling network forconnection set up and carry out other management functions.

It should further be noted that the down-converter 405 and up-converter407 contain separate RF electronics that include slight imperfections sothat they may not be perfectly matched. For this reason, atransmit/receive compensation is performed using additional compensationweights. The purpose of this compensation is to compensate for thedifferences in phase and amplitude introduced into the signals by thetransmit and receive RF electronics. By applying the compensationweights, the same beam pattern is produced on the transmit side as onthe receive side.

FIGS. 79A-79D are a schematic block diagram that depicts the overalldigital signal processing architecture layout within the base stations110. The base station 110 is laid out into a radio frequency chassisportion 2500 and a digital chassis portion 2510. The multiple elementantenna array 120, that for ease of illustration is depicted in FIGS.79A-79D as comprising four antennas, connects to correspondingtransmit/receive modules 2512. Each transmit/receive module 2512includes the transmit/receive switch 400, as well as a receiver, atransmitter, and an amplifier. It should be noted that in accordancewith one advantageous aspect of the present invention, each antennaelement is provided with an individual amplifier. By using thisdistributed amplifier configuration instead of one large amplifier topower the entire antenna array, power is saved. In addition, in theevent of amplifier failure, only one of multiple antenna elements failsrather than the entire antenna array. Thus, the present inventionprovides for graceful degradation of signal quality in the event of anamplifier failure.

An analog-to-digital converter/digital-to-analog converter pair 2515provides for analog-to-digital and digital-to-analog conversion of thereceived and transmitted signals. The digitized received signals enterthe digital chassis 2510, while the digital transmit signals areprovided as an output of the digital chassis 2510.

The digital chassis 2510 includes a G-link interface circuit thatprovides outputs to a plurality of receiver LPAs 2520 via a plurality of32-bit busses. The LPAs 2520 perform the FFTs and channel estimation inparallel (e.g., one of the LPAs performs signal processing on each ofthe even symbols, while the other performs equivalent signal processingsteps on the odd receive symbols).

The LPAs 2520 provide the processed signals to LPAs 2530, that aresubstantially similar in construction to the LPAs 2520 and the LPAs 2110and 2120. The LPAs 2530 perform QR decomposition and output thedecomposed signals to LPA cards 2540.

The LPA cards 2540 perform matrix operations involved in thenull-steering and code-nulling procedures. The retrodirective weightscalculated within the LPA cards 2540 are provided as inputs to LPA cards2550 in the transmitter path for use during data spreading, beamforming, and generating IFFTs.

An additional LPA card 2560 is provided as a digital signal processingengine for the transmitter/receiver calibration (i.e., T/Rcompensation). The T/R calibration LPA card 2560 communicates with aprobe antenna 2565 via a G-link interface, ananalog-to-digital/digital-to-analog converter, and a transmit/receivecalibration module 2570. The transmit/receive calibration module 2570includes a receiver, a transmitter, a transmitting amplifier, and atransmit/receive switch. As described briefly above, the purpose of theprobe antenna is to compensate for distortion due to the transmitter andreceiver paths through the base station 110. That is, thetransmit/receive modules 2512 introduce a certain amount of distortionand phase delay into the transmitted and received signal so that it isnecessary to compensate for these distortions to provide an accurateproduction of the transmit and receive signals. The probe antenna pathacts like a remote station so that when the base station 110 istransmitting from the antenna array 120, this information is received onthe probe 2565. Conversely, when the probe antenna 2565 is transmitting,the antenna array 120 of the base station 110 is receiving the knownsignal transmitted by the probe antenna 2565. By signal processingperformed within the transmit/receive calibration LPA card 2560, thedifferential amplitude and phase across the phase transmitter andreceiver paths can be determined. Thus, the base station 110 cancompensate for these distortions by means of the signals transmitted andreceived by the probe antenna 2565.

A global positioning system antenna 2580 receives GPS timing to providea reference clock for each of the local oscillators within the basestation 110. This ensures that accurate synchronization can be obtainedthroughout the entire wireless communication system 100.

FIGS. 6 and 7 show alternative embodiments of the directional antennaarrays 120 that may be used in the system of the present invention. Afirst embodiment of the base station antenna implementation isdesignated generally as 120 a. The antenna 120 a is a circular patchslot array antenna including a protective RADOME 505 available fromRADIX Technologies, Inc. of Mountain View, Calif., a generallycylindrical housing 507, and a support pole 510. A plurality ofmulti-element vertical patch arrays 515 are depicted in cutaway in FIG.6. Each of the patch arrays 515 are capable of directionally emittingradio frequency signals so as to provide beam forming capabilitiesnecessary for the proper implementation of the present invention. In oneembodiment, the height of the cylindrical portion 507 is approximately18″, while the diameter of the RADOME 505 is approximately 5-16″.

In one advantageous embodiment, the antenna 120 a includes a verticalstack of 4 microstrip patch antennas. Four of these stacks willrespectively be oriented to cover four 90° quadrants. Thus, a total of16 circumferential stacks of microstrip flared-notch antennas (whereeach vertical stack comprises eight notches) will be included on thebase antenna 120 a. For both the remote and base antennas, the preferredsensor element spacing is one-half wavelength.

FIG. 7 depicts a second implementation of the base station antenna ofthe present invention that is generally designated as 120 b. The antenna120 b includes a RADOME 520, a generally cylindrical portion 525, and asupport pole 530. The RADOME 520 is approximately 18-24″ in diameterwhile the cylindrical portion 525 is approximately 14″ in height. Asshown in cutaway, the antenna 120 b includes a flared circular hornconfiguration 535 as well as a plurality of monopole transmissionelements 540. The monopole elements 540 may be used for beam formingpurposes such as that that is necessary for the optimum operation of thepresent invention.

FIG. 80 is a transceiver block diagram showing the main structuralelements of the down converter 305 depicted in FIG. 72. As shown in FIG.80, the antenna 190 and the transmit/receive switch 300 connect tobandpass filters 702, 704 that, in turn, connect to amplifier 706, 708,respectively. The path through the filter 702 and the amplifier 706constitutes the receive path that is part of the down converter 305,while the path that is through the amplifier 708 and the bandpass filter704 constitutes part of the transmission pass that is a part of the upconverter circuit 307. The output of the amplifier 706 and the input ofthe amplifier 708 connect to a switch 710. The switch 710 is used toswitch between the transmission and receiving paths associated with thedown and up converters 305, 307, respectively.

Although the up converter 307 and the down converter 305 are representedin FIG. 72 as functionally distinct blocks, it will be appreciated byone of ordinary skill in the art that the same structural elements maybe used to perform the functions of both the up converter and the downconverter in an architecture that reuses amplifiers and saw filterswithin the transmitter and receiver path. The switch 710 connects to abandpass filter 712. In one advantageous embodiment, the bandpass filter712 has a bandpass frequency between 1,865 MHz and 1,950 MHz. Thebandpass filter 712 connects to a multiplier 715 that receives an inputfrom a first local oscillator having an oscillation frequency of 1667.5MHz. The multiplier 715 connects to a digital attenuator circuit 720that receives a gain control input from the demodulator 310 (see FIG.72). The digital attenuator 720 connects to an amplifier 724 via aswitching circuit 722. The switching circuit 722 allows the amplifier724 to be used bidirectionally in both the transmitter and receivepaths. That is, when switched in a first direction, the output of theamplifier 724 connects to the digital attenuator circuit 720 while whenswitched in a second mode, the input of the amplifier 724 connects tothe digital attenuation circuit 720. By using the same amplifier (i.e.,the amplifier 724) in both the transmitter and receiver paths the sameamplifier characteristics are observed in both paths so thattransmission and reception compensation is greatly simplified. Theswitching network 722 further connects to a summing circuit 725.

When operating in a receiving mode, the summing circuit 725 acts as asignal splitter while, when in the transmitting mode, the summingcircuit 725 acts to linearly add a pair of input signals. The summingcircuit 725 connects to parallel amplification and filtering pathshaving corresponding elements. Specifically, one input to the summingcircuit 725 comprises a saw bandpass filter 730 having a centerfrequency of 270 MHz and a bandwidth of 1.5 MHz. A corresponding sawbandpass filter 732 has a center frequency of 200 MHz and a bandwidth of1.5 MHz. The bandpass filters 730, 732 connect, respectively, toamplifiers 738, 740 via switching networks 734, 736. Again, theswitching networks 734, 736 insure that identical amplifiercharacteristics are observed in both the transmit and receive paths. Theamplifiers 738, 740 advantageously provide an amplification factor. Theswitching circuits 734, 736 connect to corresponding saw bandpassfilters 742, 744. The bandpass filter 742 has a center frequency ofapproximately 280 MHz and a bandwidth of 1.5 MHz, while the bandpassfilter 744 has a center frequency of 200 MHz and a bandwidth of 1.5 MHz.The bandpass filters 742, 744 connect, respectively, to correspondingamplifiers 750, 752 via switching networks 746, 748. The amplifiers 750,752 advantageously provide an amplification factor. The switchingnetworks 746, 748 connect to corresponding multipliers 754, 756. Themultiplier 754 receives a local oscillator input signal oscillating at281.25 MHz, while the multiplier 756 receives a local oscillator inputsignal oscillating at approximately 201.25 MHz.

The multipliers 754, 756 connect to corresponding low pass filters 758,760, that in turn connect to switches 762, 764, respectively. The switch762 receives an input signal from an amplifier 766 and provides anoutput signal to an amplifier 768, while the switch 764 receives aninput signal from an amplifier 770 and provides an output signal to anamplifier 772. The amplifiers 766 through 772 advantageously have anamplification factor. Amplifiers 766, 770 form a part of thetransmission path, and therefore properly belong to the up converter307, while the amplifiers 768, 772 belong to the reception path andtherefore, properly belong to the down converter 305 of FIG. 72. Theamplifiers 766, 770 connect to digital-to-analog converters 774, 778,respectively. The digital-to-analog converters 774, 778 also comprise aportion of the up converter 307 and receive a digital-to-analog clockpulse from the synchronization circuit 312. The amplifiers 768, 772connect to analog-to-digital converters 776, 780, also comprise aportion of the down converter 305 that receive analog-to-digitalconverter clock inputs from the synchronization circuit 312 (see FIG.72).

The inputs to the digital-to-analog converters 774, 778 are receivedfrom the modulation circuit 345, while the outputs of theanalog-to-digital converters 776, 780 are provided as inputs to thedemodulation circuit 310.

The operation of the up/down converter circuit depicted in FIG. 80 willfirst be described with reference to the received path and will next bedescribed with reference to the transmission path. Within the receivedmode, signals picked up by the antenna 120 are transmitted to the switch300 and passed through the bandpass filter 702 so as to attenuate anysignals that are not within the frequency band of interest (i.e.,frequencies between 1,865 MHz and 1,950 MHz). The filtered signals arethen amplified by an amplification factor within the amplifier 706. Theoutput of the amplifier 706 is provided as an input to the switch 710that allows the amplified signal to be passed through the bandpassfilter 712 that further filters out any undesired signals outside of thedesignated bandpass range.

Signals that are allowed to pass through the filter 712 are multipliedby the local oscillator frequency at 1,667.5 MHz within the multiplier715. Thus, the multiplier 715 acts as a synchronous detector that may beused to cause a first down conversion of the signal from approximatelythe 2 GHz range down to the 200 to 300 MHz range. This down-convertedsignal is then attenuated by means of the digital attenuation circuit720 and amplified with an amplification factor by means of the amplifier724. The down-converted signal is then split within the signal splitter725 so that one portion of the signal enters the saw bandpass filter 730while an identical portion of the signal enters the saw bandpass filter732.

The portion of the signal that enters the bandpass filter 730 isfiltered to attenuate signals outside of the frequency range of 279.25MHz and 280.75 MHz. This filtered signal is then amplified by a factorvia the amplifier 738 and is then filtered again through the filter 742having substantially identical characteristics to the filter 730. Onceagain, the filtered signal is amplified by the amplifier 750 with anamplification factor and this signal is input to the multiplier 754. Themultiplier 754 acts as a synchronous detector that converts the signaloutput by the amplifier 750 to substantially a base band signal bymultiplying the oscillator signal at 281.25 MHz. The base band signal isthen passes through the low pass filter 758 and from there is suppliedas an input to the amplifier 768 via the switch 762. The amplifier 768amplifies the base band signal by a factor and this signal is thenconverted to digital data by means of the analog-to-digital converter776.

The second portion of the signal output by the splitter 725 follows asubstantially similar path to that followed by the first portion of thesignal output by the splitter 725, with the exception that the secondportion of the signal is filtered to pass bandwidths between 199.25 MHzand 200.75 MHz. Furthermore, this portion of the signal is synchronouslydetected within the multiplier 756 by means of a local oscillator signalat 201.25 MHz. In this manner, signals received by the antenna 120 areasynchronously detected, down-converted to the base band level, anddigitized so as to provide digital information to be demodulated by thedemodulator 310.

The transmission path for signals that are to be transmitted by thehigh-bandwidth base station 110 is substantially the same through the upconverter as through the down converter with the exception that theorder of the signal processing steps is reversed. Specifically,modulated digital signals serve as the inputs to digital-to-analogconverters 774 and 778, so as to produce analog signals that areamplified by the amplifiers 766 and 770, respectively. The amplifiedanalog signals pass through the switching circuits 762, 764 and arefiltered by respective low pass filters 758, 760. Along the first paththe analog signal is up-converted by modulation (i.e., multiplication)with the local oscillator signal at 281.25 MHz while the second signalis up-converted by modulation with an oscillator at 201.25 MHz. Thefirst modulated signal is then amplified and filtered via the amplifiers750, 738 and the filters 742, 730 so as to provide a well defined signalbetween 200 and 79.25 MHz and 280.75 MHz. The second signal is likewiseamplified and filtered via the amplifiers 752, 740 and the filters 744,732, so as to provide a well defined signal within the frequency rangeof 199.25 MHz and 200.75 MHz. The two signals that are output from thebandpass filters 730 and 732 are provided as inputs to the summingcircuit 725. The summing circuit 725 linearly adds the two inputsignals, and these signals are amplified by the amplifier 724. Thedigital attenuation circuit 720 then attenuates the amplified outputsignal and the multiplier 715 further up converts this signal bymultiplication with the oscillator frequency at 1,667.5 MHz. In thismanner, the original input signals containing the communicationinformation are up-converted to the transmission frequency range. Thesignal to be transmitted is then filtered between 1,865 and 1,950 MHzwithin the filter 712 and the signals amplified in the amplifier 708after passing through the switch 710. The amplified transmission signalis further filtered within the bandpass filter 704 and this filtered andamplified signal is provided as an output to the antenna 120 via thetransmission/receive switch 300.

FIG. 80A is a schematic block diagram showing the main internalfunctional elements of the synchronization circuitry 312. As shown inFIG. 80A, the synchronization circuit 312 includes a frequencycontroller 785 that connects to a 40 MHz reference oscillator 787 havinga 2-bit input from a data clock (not shown). The 40 MHz referenceoscillator 787 outputs a signal to a divide-by-eight binary counter 789that, in turn, supplies the output signal references for localoscillators 791, 793 and 795. The local oscillator 791 provides theoscillation frequency at 1,667.5 MHz, while the oscillators 793, 795,respectively, provide the oscillation frequencies of 281.25 MHz and201.25 MHz. Finally, the divide-by-eight binary counter 789 furtherprovides a clock input pulse for each of the analog-to-digital anddigital-to-analog converters 774 through 780.

FIG. 81 depicts a schematic block diagram of the main elements of thedown converter 405 within the base station 110 depicted in FIG. 73.Specifically, the antenna 120 connects to a bandpass filter 802 via thetransmit/receive switch 400 while the switch 400 is in the receive mode.The filter 802 passes frequencies about 1,865 MHz and below 1,950 MHz.The filter 802 connects to the input of an amplifier 804 that, in turn,connects to a second bandpass filter 806 that has substantially the samecharacteristics as a filter 802. The filter 806 provides an input to amultiplier 809 that also receives inputs from a local oscillator (notshown in FIG. 81) at an oscillation frequency of 1,667.5 MHz. The outputof the multiplier 809 connects to a digital attenuator 811 that receivesa gain control input fee as the demodulator circuit 410 (see FIG. 73).The output of the digital attenuator 811 serves as the input to anamplifier 813 having an amplification factor.

The amplified signal output from the amplifier 813 enters a signalsplitter 815 that divides the signal into, for example, sixsubstantially identical portions. Each of the six signals output by thesplitter 815 are filtered, amplified, down-converted and digitized insubstantially the same way.

The first signal enters a bandpass filter 817 having a center frequencyof 281.5 MHz with a bandwidth of 1.5 MHz. The output of the bandpassfilter 817 serves as an input to an amplifier 819 having anamplification factor. The output of the amplifier 819 serves as theinput to a bandpass filter 821 having substantially the samecharacteristics as a bandpass filter 817. The output of the bandpassfilter 821 connects to an amplifier 823 having an amplification factor,while the output of the amplifier 823 serves as the input to amultiplier 825. The multiplier 825 also receives a local oscillatorinput at 282.5 MHz so as to act as a synchronous detection circuit thathas an output connected to a low pass filter 827. The output of the lowpass filter 827 serves as the input to an amplifier 829, while theoutput of the amplifier 829 serves as the input to an analog-to-digitalconverter 831. The analog-to-digital converter 831 further receives a 10MHz clock input from the frequency reference circuit 409 (see FIG. 73).The output of the analog-to-digital converter 831 serves as the input tothe demodulator circuit 410 in FIG. 73.

The second portion of the signal output from the signal splitter 815 isinput to a saw bandpass filter 833 having a center pass frequency of 280MHz and a bandwidth of 1.5 MHz. The bandpass filter 833 connects to theinput of an amplifier 835 that, in turn, outputs a signal to a bandpassfilter 837 having substantially the same characteristics as the bandpassfilter 833. The output of the filter 837 serves as the input to anamplifier 839 having an amplification factor. The output of theamplifier 839 connects as an input to a multiplier circuit 841 whichalso receives a local oscillator signal at 282.5 MHz. The output of themultiplier circuit 841 serves as the input to a low pass filter 843that, in turn, connects to the input of an amplifier 845 having anamplification factor. The output of the amplifier 845 is input to ananalog-to-digital converter 847 that operates off of ananalog-to-digital clock of 10 MHz. The 10 MHz clock is received from thefrequency reference circuit 409 of FIG. 73. The output of theanalog-to-digital converter 847 serves as an input to the demodulatingcircuit 410 (see FIG. 73).

The third portion of the signal output by the signal splitter 815 entersa bandpass filter 849 that has a center pass frequency of 278.5 MHz anda bandwidth of 1.5 MHz. The output of the bandpass filter 849 enters theinput of an amplifier 851 having an amplification factor, while theoutput of the amplifier 851 connects to the input of a bandpass filter853 having substantially the same bandpass characteristics as the filter849. The output of the filter 853 connects to the input of an amplifier855 that, in turn, connects to a multiplier 857 that connects to ananalog-to-digital converter 863 via a low pass filter 859 and anamplifier 861. The amplifier 855, the multiplier 857, the low passfilter 859, the amplifier 861, and the analog-to-digital converter 863are substantially identical to the corresponding elements 823, 825, 827,829 and 831, and function in substantially the same manner.

The fourth portion of the signal output from the signal splitter 815enters a bandpass filter 865 having a center bandpass frequency of 201.5MHz and a bandwidth of 1.5 MHz. The output of the bandpass filter 865serves as the input to an amplifier 866 having an output connected to abandpass filter 867 that has substantially identical filteringcharacteristics as the bandpass filter 865. The output of the bandpassfilter 867 connects to the input of an amplifier 868 having anamplification factor. The output of the amplifier 868 connects to amultiplier 869 that also receives a local oscillator frequency of 202.5MHz. Thus, the multiplier 869 acts as a synchronous detector thatoutputs a down-converted base band signal to a low pass filter 870. Thelow pass filter 870 provides an input to an amplifier 871 having anamplification factor and the output of the amplifier 871 serves as theinput to an analog-to-digital converter 872 that receives a 10 MHzanalog-to-digital converter clock from the frequency reference circuit409. The output of the analog-to-digital converter 872 serves as aninput to the demodulating circuit 410 (see FIG. 73).

The fifth and sixth portions of the signals output by the signalsplitter 815 are provided as inputs to analog-to-digital converters 880,888, respectively, via bandpass filters 873, 881, amplifiers 874, 882,bandpass filters 875, 883, amplifiers 876, 884, multipliers 877, 885,low pass filters 878, 886, and amplifiers 879, 887, respectively. Eachof the circuit elements between the splitter 815 and theanalog-to-digital converters 880, 888 are substantially identical totheir corresponding elements between the signal splitter 815 and theanalog-to-digital converter 872, with the exception that the bandpassfilters 873 and 875 have a center frequency of 200 MHz and the bandpassfilters 881, 883 have a center pass frequency of 198.5 MHz.

The operation of the down converter portion of the base station 110 issubstantially similar to that of the down converter portion of thehigh-bandwidth base station 110. Specifically, signals received by theantenna 120 and switched to the receiving path by the switch 400 arefiltered and amplified by means of the filters 802, 806 and theamplifier 804. Subsequently, the signal is down-converted to a lowerfrequency band by synchronous detection within the multiplier 809. Afterthe first down conversion step, the signal is digitally attenuated bymeans of the attenuator 811 and then amplified by means of the amplifier813. The signal is then split into a plurality of substantiallyidentical signals that each follow a different detection path. Each ofthe detection paths is substantially identical, with the exception thateach path down converts the detected signal into a different base bandfrequency range. Thus, for example, the first portion of the splitsignal is filtered about a center frequency of 281.5 MHz by the bandpassfilters 817, 821, and is amplified by the amplifiers 819, 823. Thisfiltered signal then is synchronously detected by the multiplier 825 andconverted to base band. This base band signal is subsequently filtered,amplified and digitized within the low pass filter 827, the amplifier829, and the analog-to-digital converter 831. This sequence of detectionis substantially the same for each of the six signal portions output bythe signal splitter 815, with the exception that the bandpass filtersoperate at different centering frequencies and the local oscillatorsignals that serve as inputs to the various multipliers are differentfor the bottom three signal portions than for the top three signalportions.

FIG. 81A is a simplified schematic block diagram showing the maininternal components of the frequency reference circuit 409. As shown inFIG. 81A, the frequency reference circuit 409 includes a frequencycontrol circuit 890, a 40 MHz reference oscillator 891, and adivide-by-four circuit 892. The divide-by-four circuit 892 providesoutputs to local oscillators 893, 894 and 895, as well as to each of theanalog-to-digital converter circuits and the digital-to-analog convertercircuits (see FIG. 82). The local oscillator 893 provides the 1,667.5MHz output signal, while the local oscillators 894 and 895,respectively, provide the 281.25 and the 201.25 MHz oscillator signals.

FIG. 82 is a schematic block diagram that shows the main internalcomponents of the up converter 407 along the transmission path of thebase station 110 (see FIG. 73). The antenna 120 connects to a bandpassfilter 902 via the switch 400 when the switch 400 is in the transmissionmode. The bandpass filter 902 allows frequencies between 1,865 MHz and1,950 MHz to pass. The bandpass filter 902 connects to the output of apower amplifier 904 having an amplification factor. The input of thepower amplifier 904 connects to a bandpass filter 906 having frequencypass characteristics that are substantially the same as the bandpassfilter 902. The input of the bandpass filter 906 connects to the outputof a multiplier 908 that receives a first input from a local oscillatorhaving an oscillation frequency of 1,667.5 MHz and a second input from adigital attenuator circuit 910. The digital attenuator circuit 910receives gain control inputs from the modulation circuit 445 (FIG. 73).The input of the digital attenuator circuit 910 connects to the outputof a power amplifier 912 that, in turn, receives inputs from a summingcircuit 914. The summing circuit 914 receives, in one embodiment, sixseparate inputs that are linearly added within the summing circuit 914to provide an output to the amplifier 912. Each of the six inputs to thesumming circuit 914 connects to a bandpass filter having a 1.5 MHzbandwidth. Specifically, the bandpass filters 920, 930, 940, 950, 960and 970 serve as inputs to the summing circuit 914. The bandpass filters920, 930, 940, 950, 960 and 970, respectively, have center passfrequencies of 281.5 MHz, 280 MHz, 278.5 MHz, 201.5 MHz, 200 MHz and198.5 MHz. Each of the bandpass filters 920-970, respectively, connectto the outputs of amplifiers 921-971. The amplifiers 921-971,respectively, receive inputs from bandpass filters 922-972. The bandpassfilters 922-972 have substantially the same frequency pathcharacteristics as the bandpass filters 920-970. The bandpass filters922-972 each connect to outputs of amplifier circuits 923-973.

The amplifier circuits 923-973 connect to the outputs of respectivemultipliers 924-974. The multipliers 924, 934, 944 receive localoscillator input signals at an oscillation frequency of 282.5 MHz, whilethe multipliers 954, 964, 974 receive local oscillator inputs at 202.5MHz. Each of the multipliers 924-974 connect to corresponding low passfilters 925-975. The low pass filters, in turn,.receive inputs from theoutput of respective amplifiers 926-976. Finally, each of the amplifiers926-976, respectively, receive inputs from digital-to-analog converters927-977. Each of the digital-to-analog converters 927-977 receivedigital-to-analog converter clock input signals at 10 MHz from theoutput of the divide-by-four binary counter 892 (see FIG. 81A) and alsoreceive inputs from the modulator circuit 445 shown in FIG. 73.

In operation, modulated data signals serve as inputs to thedigital-to-analog converters 927-977. The digital-to-analog converters927-977 convert the modulated digital data signals into analog signalsthat are subsequently amplified by the amplifiers 926-976, and filteredby the low pass filters 925-975. The outputs of the low pass filters925-975 enter as one input of the multipliers 924-974, respectively. Thesecond inputs of the multipliers 924-974 receive local oscillator inputsat either 282.5 MHz or 202.5 MHz. Thus, the signals output from the lowpass filters 925-975 are up-converted to a first high frequency level.The up-converted signals output by the multipliers 924-974 aresubsequently amplified and filtered by means of the amplifiers 923-973and 921-971, and the filters 922-972 and 920-970. The outputs to thefilters 920-970 serve as inputs to the summing circuit 914, thatlinearly sums each of the signals applied at the six input terminals.

The summed output of the summing circuit 914 serves as an input to thepower amplifier 912. The output of the power amplifier 912 enters thedigital attenuator circuit 910 so as to fine tune the gain controlapplied to the signal output from the signal amplifier 912, and theoutput of the digital attenuator 910 serves as the first input to themultiplier 908. The second input of the multiplier 908 is the localoscillator signal at 1,667.5 MHz. Thus, the multiplier 908 serves to upconvert the signal output from the digital attenuator 910 to thetransmission frequency of the base station 110. The output of themultiplier 908 is subsequently filtered and amplified by means of thefilters 902, 906 and the amplifier 904. Finally, the output of thefilter 902 serves as the input of the switch 400 in the transmit mode,that relays this up-converted and amplified signal to the antenna 120.

Method of Dynamically Allocating Bandwidth

The bandwidth allocation method performed by the bandwidth demandcontroller (see FIG. 74) is depicted in FIG. 83. The method begins in astart block 3300. Once the bandwidth allocation method begins,initialization functions are performed including, as shown in FIG. 83,determining if the number of antenna sensor elements has been changedsince the last use of the base station 110 or remote terminal 187, 192.For example, it may be desirable to provide a base station 110 or remoteaccess terminal 187, 192 with increased spatial resolution capability sothat the base or remote station can more accurately discriminate betweenincoming signals. In such a case, the base station 110 or remoteterminal 187, 192 would be deactivated while a new antenna is installedhaving a greater number of sensor elements that, as well known in theart, would provide a greater degree of directional discrimination orspatial division for that base station or remote. Once the installationof a new antenna is complete, then the installer reactivates the base110 or remote 187, 192 and, as indicated within a decision block 3305 atest is performed to determine if a number of antenna elements ischanged. If a number of antenna elements has changed, control passes toan activity block 3310 wherein the number of tones within a tone set isredefined (e.g., to a smaller number if the number of antenna elementsincreases) so that the matrices used to calculate the complex weightsapplied to the sensors and tones within a tone set maintain the samedimensionality. Thus, as discussed above, essentially the same SINR ispreserved while processing costs are not increased. After theinitialization such as performed within the activity block 3310, controlpasses to a decision block 3315 wherein a determination is made if a newuser is requesting bandwidth. If it was determined, however, within thedecision block 3305 that the number of antenna elements has not beenchanged, then the method passes immediately to the decision block 3315to the decision block 3305.

If it is determined within the decision block 3315, that a new user hasnot requested bandwidth through the access channel, then control passesto a subroutine block 3320 wherein the bandwidth assignments alreadyallocated within the communications link are modified, if necessary, tomaximize the SINR. Control returns from the subroutine block 3320 to thedecision block 3315 until it is determined that a new user is requestingbandwidth over the control access channel.

When a new user requests bandwidth, control passes to an activity block3325 to determine how much bandwidth is requested. As discussed above,the requested bandwidth is predicated upon the type of data that istransmitted (e.g., voice, video, data, etc.) as well as the transmittingdevice. For example, if an individual telephone unit is transmitted,then as few as 8 kilobits per second of bandwidth may be requested,while if a P−1 link connected to a PBX is requesting bandwidth, as muchas 1.544 MHz will be requested. In one embodiment, the requesting devicetransmits an initialization or identification signal that indicates tothe remote station 187, 192 the bandwidth requirements of the requesteddevice.

Once the quantity of bandwidth requested is determined within theactivity block 3325, control passes to a decision block 3330 or adetermination is made if the communications channel has sufficient freebandwidth to accommodate the requesting unit. If it is determined thatthe channel does not have sufficient free bandwidth to accommodate theoptimum bandwidth requested by the new user, then control of the methodpasses to an arbitration phase wherein a determination is first madewithin a decision block 3335 if the user can use less bandwidth. If auser cannot operate with less bandwidth than requested, then the user isdisconnected and access is denied to the communications channel asindicated within activity block 3340. However, if it is determined thatthe user can operate with less bandwidth, then the base station 110 asksthe user, via the remote station 187, 192, for a lower bandwidthrequirement, as indicated within an activity block 3345. Control thenreturns to the activity block 3325 wherein the quantity of requestedbandwidth is again determined. Of course, it will be understood, thatthe base station may present a suggested bandwidth that is allowable tothe user via the remote station 187, 192 if the user is sophisticatedenough to determine if such a suggested bandwidth would be sufficient toprovide normal operation of the user communication device.

However, if it is determined that the communication channel hassufficient free bandwidth to accommodate the requesting user, thencontrol passes from the decision block 3330 to a decision block 3350wherein a test is performed to determine if there are any free tonesets. That is, if there are any tone sets that have not yet beenallocated to other users within the region of the requesting remoteterminal 187, 192. If there are free tone sets within the region of therequesting room or terminal 187, 192, then control passes to an activityblock 3355 wherein one or more of the free tone sets is allocated to theuser for use in transmitting data from the remote associated with theuser to the base within the remote spatial cell. Control then passesfrom the activity block 3355 to an activity block 3360. If there isdetermined, however, within the decision block 3350 that there are nofree tone sets, then control passes instead to an activity block 3365wherein one or more currently used tone sets are allocated to the userfor the transmission of data between the remote 187, 192 and basestation 110. It is possible that multiple tone sets will be allocated toa requesting user if a very high bandwidth is requested by the user. Itshould also be noted here, that because the tone sets are grouped intofour approximate 1 MHz bands that when multiple tone sets are allocatedto a single user to establish a separate communications channel, thesetone sets are typically within the same 1 MHz band.

Once control passes from either the activity block 3355 or the activityblock 3365 to the activity block 3360, one or more codes (i.e.,spreading codes used to modulate the various tones within the allocatedtone set or tone sets) are allocated to the user making sure that thesame code (i.e., on the same tone set) is not used by a proximate remoteto the remote connected to the new user. In this manner, maximumfrequency and code reuse is achieved by spatially separating usershaving the same tone sets and code assignments. Of course, it will beunderstood, that due to the adapted channel equalization methoddescribed above, that the spreading codes initially assigned to theremote terminals that, on line, are typically not are well-definedcodes, but rather constitute linear adapted spreading weights tomaximize the SINR. Therefore, it is highly unlikely that a newlyallocated code will be identical to any of the spreading weightsassigned to remote terminals within the same proximity as the remoteterminal assigned to the new user. As discussed in greater detail above,the criteria for modifying spreading codes assigned to each new userrequires that the spreading weights be linearly independent to provideat least one degree of freedom for each user within a given spatial cellsite.

Control passes from the activity block 3360 to a decision block 3370wherein a determination is made if the maximum constellation size (i.e.,for any arbitrary M-ary modulation format) is sufficient to maintain therequested bandwidth given the number of tone sets and codes allocated tothe user. That is, if the newly defined communication channel toleratesa sufficiently high constellation size then the required bandwidth willbe satisfied for the requesting user. However, if the channel is notalways resistant enough to handle the necessary constellation size tomaintain the bandwidth required for operation of the new user, thenadditional codes or tone sets must be allocated to the user inaccordance with the method described. Once the tone sets, codes, andmodulation format are defined for the newly requested communicationchannel, control passes to the subroutine block 3320 wherein thebandwidth assignments are modified, as necessary, to maximize the SINR.Control then returns to the decision block 3315 and the process repeatsas described.

Alternative Embodiment of the Invention: Adaptive Beamforming for PluralDiscrete Tones Followed by Combining Resultant Signals

FIG. 84A and FIG. 84B show an alternate embodiment of the invention,where the spectral processing and the spatial processing are separated.The spatial weights are computed independently for each carrierfrequency. The spatial weights are then multiplied by spectral weightswhich are again calculated separately to produce a composite weight. Inother words, the combined spectral/spatial beamformer is broken into aseparate spectral beamformer and a separate spatial beamformer whichoperate independently. FIG. 84A shows how the received signals on theantennas through M−1 are processed by the spatial beamformer to producethe coefficients A0 through AN−1. FIG. 84A also shows how the receivedsignals on the antennas 0 through M−1 are processed by the spectralbeamformer to produce the coefficients B0 through BN−1. FIG. 84B showshow the spatial coefficients A0 are applied to tone frequency 0 and howthe result thereof is then independently operated on by the spectralcoefficients B0, with the resultant signals then being transmitted onthe antennas through M−1.

Similarly, FIG. 84B shows how the spatial coefficients AN−1 are appliedto tone frequency n−1 and how the result thereof is then independentlyoperated on by the spectral coefficients BN−1, with the resultantsignals then being transmitted on the antennas 0 through M−1. Thus, itis seen how the spatial weights are computed independently for eachcarrier frequency and then the spatial weights are multiplied byspectral weights which are calculated separately to produce a compositeweight.

In one form of this alternate embodiment, the base station and theremote unit can exchange as few as two tones, one in each of the twosub-bands. The separation of 80 MHz between the two sub-bands spreadsthe tones far enough apart so that noise bursts and interfering signalsin one sub-band do not degrade the other in the other sub-band. The twotones can be separately processed by spatial spreading and despreadingand thereafter combined to form the resultant signal. This alternateembodiment has the advantage of a simplified computation, whileretaining a reasonable immunity to noise and interference.

At the receiving station, each tone received by the multi-elementantenna array is spatially despread in a process analogous to receivebeamforming. The resultant signals are then combined. A first method ofsignal-combining is equal gain combining, where the signals are addedtogether. An alternate method of signal combining is maximal rationcombining, where the output signal is chosen from the two tones havingthe better SINR.

At the transmitting station, the alternate embodiment spatially spreadsa data signal modulated with the first tone. The spatial spreading usesspatial spreading codes in a process analogous to transmit beamforming.Separately, the alternate embodiment spatially spreads the data signalmodulated with the second tone. Then, the two spatially spread signalsare combined and transmitted from the multi-element antenna array,forming a transmitted spread signal that is spectrally and spatiallyspread.

The alternate embodiment of the invention can have the spatialdespreading steps adaptively position spatial directions of the receiversensitivity towards a desired signal source and/or diminish the receiversensitivity from interfering sources. The alternate embodiment can alsohave et spreading steps adaptively position transmitted signal energy ofthe transmitted despread signal towards a source of the received spreadsignal and/or adaptively diminish the transmitted signal energy towardsinterferers. The alternate embodiment works well within the TDDprotocol.

FIG. 85A, consisting of FIGS. 85A-L and 85A-R, is a flow diagram of apreferred embodiment, describing the computational steps performed inthe base station. In the transmission portion of the base station,traffic symbols are input on line 5 to the smear matrix step 10. Linkmaintenance pilot signals are input on line 7 to the digital signalprocessor (DSP) data processing RAM 12. Stored pilot signals are outputfrom the RAM 12 to the link maintenance pilot (*LMP) register 14 and arethen applied as one input to the smear step 10. The smear matrix 16 isalso applied to the smear step 10. The output of the smear matrix 16 isalso applied to the smear step 10. The output of the smear step 10 isapplied to the gain emphasis step 20. The values from the gain RAM25 areapplied to the gain emphasis step 20 to provide output values which arethen applied to the beam form spreading step 30. Spreading weights in aspread weight RAM 32 are applied to the beam form spread step. The Xvector is output on line 40 from the beam form spread step and is sentto the transmitter for transmission to the remote station.

On the receive side of the signal processing in the base station, the Xvector from the receivers is input on line 50 to the beam form despreadstep 60. The despreading weight RAM 62 applies the despreading weightsto the beam form despread step 60. The signal output from the beam formdespread 60 is then applied to the gain emphasis step 70. Values fromthe gain RAM 25 are applied to the gain emphasis step 70. Values formthe gain RAM 25 are applied to the gain emphasis step 70. Values outputfrom the gain emphasis step 70 are applied to the desmear step 80.Values for pilot signals from the gain emphasis step 70 are stored inthe LMP register 72 and are applied to the desmear step 80. The desmearmatrix is also applied form step 74 to the desmear step 80. Trafficsymbols output from the desmear step 80 on line 82 are then available tobe utilized and further distributed in the base station. The pilotsignals output form the LMP register 72 are stored in the LMP digitalsignal processing DP RAM 76 and are then output on line 78.

Various values used in the spreading and despreading computations areupdated as is shown in FIG. 85A. The X vector input on line 50 isapplied to the updated weight step 54. The X vector input online 50 isalso applied to the data correction step 93 whose output is applied tothe update weight step 54. The updated weight values output from theupdated weight step 54 are sent to the valid weights step 56 and arethen output to the despread RAM 62. The traffic establishment support 86provides values to the property map 84 which processes traffic signalsfrom line 82 and applies the output to the smear step 89. Maintenancepilot signals online 81 are applied to the digital signal processing DPRAM 83 whose output is applied to the LMP register 85 whose output isapplied to the smear step 89. The smear matrix 87 is also applied to thesmear step 89. The output of the smear step 89 is applied to the gainde-emphasis step 91 whose output is applied to the data correlation step93 whose output is applied to the updated weight step 54 as previouslydescribed. In addition, the output from the smear step 89 is applied tothe element-wise gain covariance step 64. Another input to theelement-wise gain covariance step 64 is applied from the output of thebeam form despread step 60. The output of the element-wise gaincovariance step 64 is applied to the block normalization of elementsstep 66 which is in turn applied to the element-wise conjugation step 68which outputs the values to the gain RAM 25. In this manner, the basestation can perform both despreading operations for received signalvectors online 50 and spreading operations to transmit traffic symbolsinput on line 5, in accordance with the invention.

FIG. 85B, consisting of FIGS. 85B-L and 85B-R, shows the processing ofthe common access channel signals. Two common access signals (CAC)signals from the transmitter are processed. A first signal is processedbeing received on the input line 102 and is applied to the RMGSauto-correlation step 104, whose output goes to the digital signalmatrix step 106 whose output goes to the digital signal processor. Thecommon access channel signal online 102 is also applied to the selectungated packets step 108 and to the select gated packets step 110. Theoutput of the select ungated packets 108 is applied to the subtracteven/odd packets step 112. The output of the selected gate packets 110is applied to the apply code key step 114. The CAC code key step 116applies it's value to the apply code key step 114. the output of theapply code key step 114. The output of the apply code key step 114 isalso applied to the subtract even/odd packets step 112. The output ofthe subtract even/odd packets step 112 ids applied to the RMGSautocorrelation step 118, whose output is also applied to the compute Tmatrix step 106. The output of the compute T matrix step 106 is thenapplied to the digital signal processor.

A second one of the two CAC signals input from the receiver on line 120is applied to the select ungated packets step 122 and the select gatedpackets step 124. The output of the select ungated packets step 122 isapplied as one input to the combined gated/ungated packets step 126. Theoutput of the selected gated packet steep 124 is applied to the applycode key step 128 which also receives a signal from the CAC code keystep 130. The output of the applied code key step 128 is the secondinput to the combined gated/ungated packets step 126. The output of thecombined gated/ungated packets step 126 is applied to the apply despreadweight step 132. A signal from the digital signal processor is appliedto the rotated weight RAM 134 whose output is applied to the computedespread weight step 136. The output of the compute despread weight step136 is applied to the apply despread weight 132, whose output is sent tothe digital signal processor. In this manner, the steps shown in FIG.85B carry out processing of the common access channel signals.

Although the preferred embodiments of the invention have been describedin detail above, it will be apparent to those of ordinary skill in theart that obvious modifications may be made to the invention withoutdeparting from its spirit or essence. For example, signal constellationformats other than PSK, BPSK and QAM could be used in accordance withthe system of the present invention. Furthermore, the system couldoptimize for bit error rate (BER) rather than SINR. Also, the number oftones in a tone set, and the number of tone sets and cluster sets in aband could be selected based upon the specific application. The selectedfrequency bands could also be varied as called for by specificconditions. The TDD format could be altered based upon the multipathenvironment to insure that an effectively static channel is observed insuccessive TDD frames. The maximization of the SINR could be performedbased upon some signal property other than constant modulus, etc.Consequently, the preceding description should be taken as illustrativeand not restrictive, and the scope of the invention should be determinedin view of the following claims.

APPENDIX

In each of the following five appendices, the term “Discrete MultitoneSpread Spectrum” and the acronym “DMT-SS” refer to the term “DiscreteMultitone Stacked Carrier” and the acronym “DMT-SC” discussed above.

APPENDIX A—IMPROVED NETWORK ACCESS METHOD FOR A WIRELESS DISCRETEMULTITONE SPREAD SPECTRUM COMMUNICATIONS SYSTEM (4342)

APPENDIX B—PRIORITY MESSAGING METHOD FOR A DISCRETE MULTITONE SPREADSPECTRUM COMMUNICATIONS SYSTEM (4343)

APPENDIX C—METHOD OF POLLING REMOTE STATIONS FOR FUNCTIONAL QUALITY ANDMAINTENANCE DATA IN A DISCRETE MULTITONE SPREAD SPECTRUM COMMUNICATIONSSYSTEM (4348)

APPENDIX D—POWER MANAGEMENT IN A DISCRETE MULTITONE SPREAD SPECTRUMCOMMUNICATIONS SYSTEM (4382)

APPENDIX E—NETWORK DIRECTIVITY IN A DISCRETE MULTITONE SPREAD SPECTRUMCOMMUNICATIONS SYSTEM

IMPROVED NETWORK ACCESS METHOD FOR A WIRELESS DISCRETE MULTITONE SPREADSPECTRUM COMMUNICATIONS SYSTEM (2455-4342) BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to communications systems and methods ofoperation. More particularly, the invention relates to wireless discretemultitone spread spectrum communications systems and method ofoperation.

2. Background Discussion

Wireless communications systems, such as cellular and personalcommunications systems, operate over limited spectral bandwidth. Suchsystems must make highly efficient use of the limited bandwidth resourceto provide good service to a large population of wireless users. CodeDivision Multiple Access (CDMA) protocol has been used by wirelesscommunications systems to efficiently make use of limited bandwidth. Theprotocol uses a unique code to distinguish each user's data signal fromother users' data signals. Knowledge of the unique code with which anyspecific information is transmitted, permits the separation andreconstruction of each user's message at the receiving end of thecommunication channel.

The Personal Wireless Access Network (PWAN) described in the referencedAlamouti, et al. patent application, uses a form of the CDMA protocolknown as Discrete Multitone Spread Spectrum (DMT-SS) to provideefficient communications between a base station and a plurality ofremote units. In this protocol, the user's data signal is modulated by aset of weighted discrete frequencies or tones. The weights are spreadingcodes that distributed the data signal over many discrete tones coveringa broad range of frequencies. The weights are complex numbers with thereal component acting to modulate the amplitude of a tone while complexcomponent acts to modulate the phase of the tone. Each tone in theweighted tone set bears the same data signal. Plural users at thetransmitting station can use the same tone set to transmit their data,but each of the users sharing the tone set has a different set ofspreading codes. The weighted tone set for a particular user istransmitted to the receiving station where it is processed withdespreading codes related to the user's spreading codes, to recover theuser's data signal. For each of the spatially separated antennas at thereceiver, the received multitone signals are transformed from timedomain signals to frequency domain signals. Despreading weights areassigned to each frequency component of the signals received by eachantenna element. The values of the despreading weights are combined withthe received signals to obtain an optimized approximation of individualtransmitted signals characterized by a particular multitone set andtransmitting location. The PWAN system has a total of 2560 discretetones (carriers) equally spaced in 8 MHz of available bandwidth in therange of 1850 to 1990 MHz. The spacings between the tones is 3.125 KHz.The total set of tones are numbered consecutively from 0 to 2559starting from the lowest frequency tone. The tones are used to carrytraffic messages and overhead messages between the base station and theplurality of remote units. The traffic tones are divided into 320traffic partitions, with each traffic channel requiring at least onetraffic partition of 72 tones.

In addition, the PWAN system uses overhead tones to establishsynchronization and to pass control information between the base stationand the remote units. A Common Link Channel (CLC) is used by the basestation to transmit control information to. the remote units. A CommonAccess Channel (CAC) is used to transmit messages from the remote unitto the base station. There is one grouping of tones assigned to eachchannel. These overhead channels are used in common by all of the remoteunits which they are exchanging control messages with the base station.In the PWAN system, Time Division Duplexing (TDD) is used by the basestation and the remote unit to transmit data and control information inboth directions over the same multitone frequency channel. Transmissionfrom the base station to the remote unit is called forward transmissionand transmission from the remote unit to the base station is calledreverse transmission. The time between recurrent transmissions fromeither the remote unit or the base station is the TDD period. In everyTDD period, there are four consecutive transmissions in each direction.Data is transmitted in each burst using multiple tones. The base stationand remote unit must synchronize and conform to the TDD timing structureand both the base station and the remote units must synchronize to aframing structure. All remote units and base stations must besynchronized so that all remote units transmit at the same time and thenall base stations transmit at the same time. When a remote unitinitially powers up, synchronization is acquired from the base stationso that it can exchange control and traffic messages within theprescribed TDD time format. The remote unit must also acquire frequencyand phase synchronization for the DMT-SS signals so that the remote isoperating at the same frequency and phase as the base station.

In existing wireless systems, a traffic link is established between theremote unit and the base station before a set up message is sent to thenetwork switch to establish a call in a Public Switch Telephone Network(PSTN). The establishment of the remote unit traffic link is timeconsuming and uses valuable bandwidth while the user “waits” for theconnection to be established. Also, until a set up message is sent tothe network switch, the remote user does not receive a dial tone unlikewireline systems in which the user receives a dial tone as soon as thetelephone goes “off hook”. Accordingly, there is a need in spreadspectrum wireless systems to eliminate “waiting periods” therebyconserving bandwidth for other users. There is a further need to providea remote user with dial tone as soon as the remote unit goes “off hook”which duplicates the same level of service as in wireline systems.

SUMMARY OF THE INVENTION

A wireless discrete multitone spread spectrum communications method andsystem are disclosed which minimizes call set up time in establishing aconnection over a wireless link between a remote user and a base stationin connecting the remote user to a network switch in a communicationsystem, e.g., PSTN. By minimizing the call set up time, valuablebandwidth is made available to other users of the system. In the system,a remote station serves multiple subscribers, each subscriber coupledthrough a transmitter/receiver to a base station over a wireless link.The wireless link includes a Common Link Channel (CLC) for controlmessages and a Common Access Channel (CAC) for traffic. Messages areinterchanged between the remote station and the base station using PWAN.The base station is coupled to a network switch in a communicationnetwork, e.g., PSTN. To minimize call set up time, the remote stationtransmits an initial set up request message to the switch over the CACbefore a traffic channel is established between the base station and theremote station, when a subscriber telephone goes “off hook”. The set uprequest message includes the remote station ID and the subscriber linenumber. In response to the set up request message, the base stationaccesses a database to identify the particular subscriber and relatedprofile for use by the network switch. Simultaneously, the base stationinitiates a wireless traffic connection between the remote user and thebase station and activates a set up processor which sends the setuprequest message to the network switch for processing. The network switchprovides a dial tone to the subscriber at the remote station afterreceipt of the set up request message. In the event the base station isunable to establish a traffic channel between the remote user and thebase station due to radio propagation or other problems, an errorprocessor is activated by the base station. The error processororiginates a signal or instruction to the network switch to disassembleor “tear down” the set up connection. By sending the set up message tothe network switch before a traffic channel is established, call set uptime is minimized; bandwidth is conserved; and the remote user receivesdial tone equivalent to wireline communication.

BRIEF DESCRIPTION OF THE DRAWING

FIG. 86 is a block diagram of a plurality of remote stations coupled toa base station over a wireless link using discrete multitone spreadspectrum communication and incorporating the principles of the presentinvention.

FIG. 87 is a block diagram of a base station included in FIG. 86.

FIG. 88 is a flow diagram which implements the operation of theinvention of FIGS. 86 and 87.

DESCRIPTION OF PREFERRED EMBODIMENT

In FIG. 86, a remote station “X” and a remote station “Y” are coupled toa base station “Z” over a wireless link using traffic channels for datatraffic, and a common access channel (CAC) and a common link channel(CLC) for control information. Each remote station includes a pluralityof subscribers coupled to a transmitter/receiver which uses the discretemultitone spread spectrum protocol for transmissions. Communicationbetween the remote stations and the base station is performed in themanner described in the above cited S. Alamouti et al. and E. Hoole etal. applications.

The base station includes receiver/transmitters for each channel coupledto the remote stations. The base station further includes databases forsubscriber spreading weights and despreading weights used in processingthe messages between the remote stations and the base station. A callset up processor in the base station is used in transmitting call setrequest messages to the network switch. An error processor is activatedby the base station when a traffic link can not be established betweenthe base and a remote station, as will be explained hereinafter. Lastly,the base station is coupled a wireline link to a network switch servinga public switch telephone network or the like.

The personal wireless access network (PWAN) system described in thereferenced Alamouti, et al. patent applications provides a more detaileddescription of the base station. The base station transmits informationto multiple remote stations in its cell. The transmission formats arefor a 64 kbits/sec traffic channel, together with a 4 kbps link controlchannel (LCC) between the base and a remote station. The binary sourcedelivers data to the sender's transmitter at 64 kbits/sec. Thistranslates to 48 bits in one transmission burst. The information bitsare encrypted according to a triple data encryption standard (DES)algorithm. The encrypted bits are then randomized in a datarandomization block. A bit to octal conversion block converts therandomized binary sequence into a sequence of 3-bit symbols. The symbolsequence is converted into 16 symbol vectors. The term vector generallyrefers to a column vector which is generally complex. One symbol fromthe LCC is added to form a vector of 17 symbols.

The 17-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the LCCsymbol). This process employs convolutional encoding that converts theinput symbol (an integer between 0 and 7) to another symbol (between 0and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is signal withinthe set of 16QAM (or 16PSK) constellation signals. (The term signal willgenerally refer to a signal constellation point.)

A link maintenance pilot signal (LMP) is added to form an 18-signalvector, with the LMP as the first elements of the vector. The resulting(18×1) vector is pre-multiplied by a (18×18) forward smearing matrix toyield a (18×1) vector b.

Vector b is element-wise multiplied by the (18×1) gain preemphasisvector to yield another (18×1) vector, c, where p denotes the trafficchannel index and is an integer. Vector c is post-multiplied by a (1×32)forward spatial and spectral spreading vector to yield a (18×32) matrixR(p). The number 32 results from multiplying the spectral spreadingfactor 4 and spatial spreading factor 8. The 18×32 matricescorresponding to all traffic channels carried (on the same trafficpartition) are then combined (added) to produce the resulting 18×32matrix S.

The matrix S is partitioned (by groups of four columns) into eight(18×4) submatrices (A₀ to A₇). (The indices 0 to 7, corresponds to theantenna elements over which these symbols will eventually betransmitted.) Each submatrix is mapped to tones within one trafficpartition.

A lower physical layer places the baseband signals in discrete Fouriertransfer (DFT) frequency bins where the data is converted into the timedomain and sent to its corresponding antenna elements (0 to 7) fortransmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next forward transmission burst.

In FIG. 87, a base station further includes a spectral and spatialdespreading processor 312 which interacts with the spreading anddespreading databases in accordance with the S. Alamouti et al.application previously cited. The processor is coupled to a decoderwhich provides an output to a vector disassembly buffer 316 forgenerating subscriber data originated in a call. The decoder is alsocoupled to a subscriber database buffer which contains informationrelated to the subscriber name, number and other standard subscriberinformation including, for example, subscriber profiles. The output ofthe database buffer is provided to a call set up processor 330 or anerror processor 322 as will be described in more detail hereinafter. Theprocessors 330 and 322 are connected to the network switch 202.

The operation of FIGS. 86 and 87 will now be described in conjunctionwith FIG. 88. In step 710, a subscriber coupled to a remote stationoriginates a call which initiates an “off hook” condition at thestation. A set up connection request is initiated by the remote stationin a step 720. The remote station transmits setup request message; theremote station ID and subscriber line number to the base station using aCAC tone. The base station responds to the set up connection request instep 730 and accesses the database 320 to identify the subscriber andobtain the subscriber profile. Simultaneously, in steps 740 and 743, thebase station initiates the establishment of a traffic channel to theremote station and sends the set up request; remote user ID, subscriberline number and subscriber profile to the network switch 202. Thenetwork switch initiates the set up in a step 745 and provides dial toneto the subscriber at the remote station. During the process ofestablishing the traffic channel, the base station performs a test in astep 742 to determine whether a traffic channel has been establishedbetween the remote station and the base station. In some instances, theradio propagation characteristics of the channel are such that a linkcannot be established. In the event that a link cannot be established, a“no” condition from the test 742 activates the error processor 322 whichprovides the network switch in a step 744 with a signal to a logicdevice which signals the network switch to disassemble or “tear down”the call set up in the PSTN, if the base station has sent the set uprequest message. In response to the logic device signal, the networkswitch in a step 749 “tears down” the PSTN connections and the processends. In the instance where the traffic channel is completed, a “yes”condition sends a signal to a logic device whereupon the network switchcompletes the call in a step 747, provided the call setup has beeninitiated and dial tone provided to the subscriber by the networkswitch.

While our invention has been described in terms of the specificembodiment, various modifications can be made without departing from thespirit and scope of the invention as defined in the appended claims.

TITLE OF THE INVENTION: “PRIORITY MESSAGING METHOD FOR A DISCRETEMULTITONE SPREAD SPECTRUM COMMUNICATIONS SYSTEM”2455/4343 BACKGROUND OFTHE INVENTION

1. Field of the Invention

This invention involves improvements to communications systems andmethods in a wireless discrete multitone spread spectrum communicationssystem.

2. Description of Related Art

Wireless communications systems, such as cellular and personalcommunications systems, operate over limited spectral bandwidths. Theymust make highly efficient use of the scarce bandwidth resource toprovide good service to a large population of users. Code DivisionMultiple Access (CDMA) protocol has been used by wireless communicationssystems to efficiently make use of limited bandwidths. The protocol usesa unique code to distinguish each user's data signal from other users'data signals. Knowledge of the unique code with which any specificinformation is transmitted, permits the separation and reconstruction ofeach user's message at the receiving end of the communication channel.

Adaptive beamforming technology has become a promising technology forwireless service providers to offer large coverage, high capacity, andhigh quality service. Based on this technology, a wireless communicationsystem can improve its coverage capability, system capacity, andperformance significantly.

The personal wireless access network (PWAN) system described in thereferenced Alamouti, Stolarz, et al. patent application, uses adaptivebeamforming combined with a form of the CDMA protocol known as discretemultitone spread spectrum (DMT-SS) to provide efficient communicationsbetween a base station and a plurality of remote units. The PWAN systemavoids loading normal, high priority traffic channels with systemmanagement information. Instead, system management information is sentas messages on the link control channel. There are two types of systemmanagement messages. The first are system management messages that arerelatively important but which are not time critical, such as softwaredownloads. The second are system management messages that are timecritical, such as call control messages, connect messages,acknowledgements for call control, and signaling. The link controlchannel is the only channel available for sending all of these systemmanagement messages. What is needed is a way to insure that timecritical system management messages are given priority over those thatare not time critical.

SUMMARY OF THE INVENTION

The invention disclosed herein is a new method to make the mostefficient use of the scarce spectral bandwidth in a wireless discretemultitone spread spectrum communications system.

The invention manages the exchange of system management messages overthe link control channel between a remote station and the base stationso that time critical system management messages are given priority overthose that are not time critical. The invention applies either to theremote station sending system management messages to the base station orto the base station sending system management messages to the remotestation.

The following summarizes operation of the remote station sending systemmanagement messages to the base station. The remote station and the basestation are part of a wireless discrete multitone spread spectrumcommunications system. The remote station, which is the sending stationin this example, includes a priority message processor that selects theorder in which system management messages are transmitted over the linkcontrol channel. The order of selection is by the time criticality ofthe message. Those messages having a greater time criticality areselected to be transmitted first. The priority message processor in thesending station is programmed to rank call control messages, connectmessages, acknowledgement messages for call control, and signalingmessages, for example, to have a greater time criticality than systemstatus messages or software downloads have. The burst size transmittedfrom a sending station is a fixed number of bits long, for exampleforty-eight bits in length. If the message to be sent is longer than theburst size, then the priority message processor at the sending stationbreaks the message into segments. In accordance with the invention, apriority interrupt flag of one bit in length is included with eachmessage segment, to identify whether the segment is the first occurringsegment in a message. This enables the sending station and the receivingstation to cooperate in managing the communication of system managementmessages having differing time criticality.

As an example, the remote station is in the middle of sending a first,multiple segment message over the link control channel to the basestation. The first message can be a status message having a low timecriticality, as assessed by the priority message processor at the remotestation. The first segment in the first message has its priorityinterrupt flag set to one, indicating that it is the first segment. Theremaining segments in the message have their respective priorityinterrupt flags set to zero, indicating that they are not the firstsegment. At an instant when a segment with a flag set to zero isscheduled by the priority message processor to be sent, the remotestation has a call control message signal, such as an off-hook signal,input to it from a local subscriber. The priority message processor atthe remote station determines that a call control message has a highertime criticality than does the currently transmitting status message. Inaccordance with the invention, the priority message processor at theremote station truncates the first message. The priority messageprocessor breaks the second message into burst sized segments that eachinclude a priority field for each segment. The priority messageprocessor assigns a priority interrupt flag value of one to the firstsegment and initiates its transmission to the base station over the linkcontrol channel. The priority message processor assigns a priorityinterrupt flag value of zero to the remaining segments of the secondmessage and buffers them for transmission over the link control channelin later bursts. The remote station then transmits a burst with a firstdiscrete multitone spread spectrum signal comprising a data trafficsignal having a data portion spread over a plurality of discrete trafficfrequencies. The remote station also transmits in the burst, a seconddiscrete multitone spread spectrum signal comprising a message segmentsignal having the first message segment of the call control message andthe priority interrupt flag portion spread over a plurality of linkcontrol channel frequencies.

In accordance with the invention, the base station receives the burstwith the first spread signal and the second spread signal. The basestation adaptively despreads the first spread signal received by usingdespreading weights, recovering the data portion. The base station alsoadaptively despreads the second spread signal received by usingdespreading weights, recovering the message segment portion and thepriority interrupt flag portion. The base station includes a prioritymessage processor that receives from the link control channel, the firstmessage segment of the call control message. The priority messageprocessor at the base station then resets a message segment buffer inthe base station and stores the first segment of the call controlmessage in the buffer, if the priority interrupt flag has a first valueof one. The first value of one for the priority interrupt flagcorresponds to a time critical message segment. This operationeffectively substitutes the more time critical call control message forthe first, status message at the base station.

When the remaining segments of the call control message are received bythe base station's priority message processor over the link controlchannel, the priority message processor at the base station concatenatesthe remaining message segments with the first received message segment,since the priority interrupt flag for those segments has a second valueof zero. The second value of zero for the priority interrupt flagcorresponds to a message segment that is not the first segment in amessage having plural segments.

In an alternate embodiment of the invention, the base stationselectively reassigns its message processing capacity from low prioritymessages it is transmitting, to more time critical messages that itreceives on the link control channel. In accordance with the invention,the base station is currently transmitting a transmitted spread signalcomprising an outgoing data traffic signal spread over a plurality ofdiscrete traffic frequencies and an outgoing message segment signalspread over a plurality of link control frequencies. This takes placeduring the transmit interval of a time division duplex session with theremote station. The outgoing message segment signal from the basestation is part of a low priority message, such as a software downloadto the remote station. During the next receive interval of the timedivision duplex session, the base station is receiving a spread signalcomprising an incoming data traffic signal spread over a plurality ofdiscrete traffic frequencies and an incoming message segment signalspread over a plurality of link control frequencies. The base stationadaptively despreads the signals received at the base station by usingdespreading weights. Then the base station detects a priority interruptflag value in the message segment signal. In accordance with thealternate embodiment of the invention, the base station acts to reassignits message processing capacity by interrupting the next scheduledtransmission of the second outgoing message segment signal. The basestation also resets the message segment buffer in the base station, andstores the incoming message segment signal therein. These steps aretaken by the base station if the priority interrupt flag has a firstvalue one. The first value of one for the priority interrupt flagcorresponds to a time critical message segment. Alternately, the basestation concatenates the incoming message segment signal with apreviously received message segment, if the priority interrupt flag hasa second value of zero. The second value of zero for the priorityinterrupt flag corresponds to a message segment that is not a firstmessage segment in a message having plural segments. In this manner, theinvention manages the exchange of system management messages over thelink control channel between a remote station and the base station sothat time critical system management messages are given priority overthose that are not time critical.

Currently, the invention has advantageous applications in the field ofwireless communications, such as cellular communications or personalcommunications, where bandwidth is scarce compared to the number of theusers and their needs. Such applications may be effected in mobile,fixed, or minimally mobile systems. However, the invention may beadvantageously applied to other, non-wireless, communications systems aswell.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 89 is an architectural diagram of the PWAN system, including remotestations transmitting to a base station.

FIG. 90 is an architectural diagram of the remote station X as a sender.

FIG. 91 is an architectural diagram of the base station Z as a receiver.

FIG. 92 is a more detailed architectural diagram of the priority messageprocessor 204 at the sending station.

FIG. 93 is a flow diagram showing the remote station as the sender andthe base station as the receiver.

FIG. 94 is a more detailed architectural diagram of the priority messageprocessor 320 at the receiving station.

DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 89 is an architectural diagram of the personal wireless accessnetwork (PWAN) system described in the referenced Alamouti, Stolarz, etal. patent application. Two users, Alice and Bob, are located at theremote station X and wish to transmit their respective data messages tothe base station Z. Station X is positioned to be equidistant from theantenna elements A, B, C, and D of the base station Z. Two other users,Chuck and Dave, are located at the remote station Y and also wish totransmit their respective data messages to the base station Z. Station Yis geographically remote from Station X and is not equidistant from theantenna elements A, B, C, and D of the base station Z. The remotestations X and Y and the base station Z use the form of the CDMAprotocol known as discrete multitone spread spectrum (DMT-SS) to provideefficient communications between the base station and the plurality ofremote station units. This protocol is designated in FIG. B1 asmulti-tone CDMA. In this protocol, the user's data signal is modulatedby a set of weighted discrete frequencies or tones. The weights arespreading weights that distribute the data signal over many discretetones covering a broad range of frequencies. The weights are complexnumbers with the real component acting to modulate the amplitude of atone while the complex component of the weight acts to modulate thephase of the same tone. Each tone in the weighted tone set bears thesame data signal. Plural users at the transmitting station can use thesame tone set to transmit their data, but each of the users sharing thetone set has a different set of spreading weights. The weighted tone setfor a particular user is transmitted to the receiving station where itis processed with despreading weights related to the user's spreadingweights, to recover the user's data signal. For each of the spatiallyseparated antennas at the receiver, the received multitone signals aretransformed from time domain signals to frequency domain signals.Despreading weights are assigned to each frequency component of thesignals received by each antenna element. The values of the despreadingweights are combined with the received signals to obtain an optimizedapproximation of individual transmitted signals characterized by aparticular multitone set and transmitting location.

The PWAN system has a total of 2560 discrete tones (carriers) equallyspaced in 8 MHZ of available bandwidth in the range of 1850 to 1990 MHZ.The spacing between the tones is 3.125 kHz. The total set of tones arenumbered consecutively form 0 to 2559 starting from the lowest frequencytone. The tones are used to carry traffic messages and overhead messagesbetween the base station and the plurality of remote units. The traffictones are divided into 32 traffic partitions, with each traffic channelrequiring at least one traffic partition of 72 tones.

In addition, the PWAN system uses overhead tones to establishsynchronization and to pass control information between the base stationand the remote units. A Common Link Channel (CLC) is used by the base totransmit control information to the Remote Units. A Common AccessChannel (CAC) is used to transmit messages from the Remote Unit to theBase. There is one grouping of tones assigned to each channel. Theseoverhead channels are used in common by all of the remote units whenthey are exchanging control messages with the base station.

In the PWAN system, Time Division Duplexing (TDD) is used by the basestation and the remote unit to transmit data and control information inboth directions over the same multi-tone frequency channel. Transmissionfrom the base station to the remote unit is called forward transmissionand transmission from the remote unit to the base station is calledreverse transmission. The time between recurrent transmissions fromeither the remote unit or the base station is the TDD period. In everyTDD period, there-are four consecutive transmission bursts in eachdirection. Data is transmitted in each burst using multiple tones. Thebase station and each remote unit must synchronize and conform to theTDD timing structure and both the base station and the remote unit mustsynchronize to a framing structure. All remote units and base stationsmust be synchronized so that all remote units transmit at the same timeand then all base stations transmit at the same time. When a remote unitinitially powers up, it acquires synchronization from the base stationso that it can exchange control and traffic messages within theprescribed TDD time format. The remote unit must also acquire frequencyand phase synchronization for the DMT-SS signals so that the remote isoperating at the same frequency and phase as the base station.

Selected tones within each tone set are designated as pilots distributedthroughout the frequency band. Pilot tones carry known data patternsthat enable an accurate channel estimation. The series of pilot tones,having known amplitudes and phases, have a known level and are spacedapart by approximately 30 KHz to provide an accurate representation ofthe channel response (i.e., the amplitude and phase distortionintroduced by the communication channel characteristics) over the entiretransmission band.

In accordance with the invention, a new method makes the most efficientuse of the scarce spectral bandwidth in a wireless discrete multitonespread spectrum communications system. The invention manages theexchange of system management messages over the link control channelbetween a remote station and the base station so that time criticalsystem management messages are given priority over those that are nottime critical. The invention applies either to the remote stationsending system management messages to the base station or to the basestation sending system management messages to the remote station.

The following describes the operation of the remote station X in sendingsystem management messages to the base station Z. The remote station andthe base station are part of a wireless discrete multitone spreadspectrum communications system. The remote station, which is the sendingstation in this example, includes a priority message processor 204 shownin FIG. 90 and in FIG. 92, that selects the order in which systemmanagement messages are transmitted over the link control channel (LCC).The order of selection is by the time criticality of the message. Thosemessages having a greater time criticality are selected to betransmitted first. The priority message processor 204 in the sendingstation is programmed by program 400 of FIG. 92, to rank call controlmessages, connect messages, acknowledgement messages for call control,and signaling messages, for example, to have a greater time criticalitythan system status messages or software downloads have. The burst sizetransmitted from a sending station is a fixed number of bits long, forexample forty-eight bits in length. If the message to be sent is longerthan the burst size, then the priority message processor 204 at thesending station uses the priority message buffer 420 in FIG. 92, tobreak the message into segments. In accordance with the invention, apriority interrupt flag “P” of one bit in length is included with eachmessage segment, to identify whether the segment is the first occurringsegment in a message. A first segment of a message, with a priorityinterrupt flag bit P=1, will be sent in a first occurring transmit bursttime. The remaining segments that are not the first segment of amessage, those segments with a priority interrupt flag bit P=0, will besent in a later occurring transmit burst times. This enables the sendingstation and the receiving station to cooperate in managing thecommunication of system management messages having differing timecriticality.

As an example, the remote station X is in the middle of sending a first,multiple segment message over the link control channel to the basestation Z. The first message can be a status message having a low timecriticality, as assessed by the priority message processor 204 at theremote station. The first segment in the first message has its priorityinterrupt flag “P” set to one, indicating that it is the first segment.The remaining segments in the message have their respective priorityinterrupt flags “P” set to zero, indicating that they are not the firstsegment. At an instant when a segment with a flag “P” set to zero isscheduled by the priority message processor 204 to be sent, the remotestation X has a call control message signal, such as an off-hook signal,input to it from a local subscriber, Alice. The priority messageprocessor 204 at the remote station X determines that a call controlmessage has a higher time criticality than does the currentlytransmitting status message. In accordance with the invention, thepriority message processor 204 at the remote station truncates the firstmessage. The priority message processor breaks the second message intoburst sized segments that each include a priority field for eachsegment. The priority message processor 204 assigns a priority interruptflag value of P=1 (one) to the first segment and initiates itstransmission to the base station Z over the link control channel. Thepriority message processor 204 assigns a priority interrupt flag valueof P=0 (zero) to the remaining segments of the second message andbuffers them in the priority message buffer 420 for transmission overthe link control channel in later bursts. The remote station X thentransmits a burst with a first discrete multitone spread spectrum signalcomprising a data traffic signal having a data portion spread over aplurality of discrete traffic frequencies. The remote station X alsotransmits in the burst, a second discrete multitone spread spectrumsignal comprising a message segment signal having the first messagesegment of the call control message and the priority interrupt flagportion “P” spread over a plurality of link control channel frequencies.

In accordance with the invention, the base station Z of FIG. 91 receivesthe burst with the first spread signal and the second spread signal. Thebase station adaptively despreads the first spread signal received byusing despreading weights in the spectral and spatial despreadingprocessor 312, recovering the data portion. The base station alsoadaptively despreads the second spread signal received by usingdespreading weights, recovering the message segment portion and thepriority interrupt flag portion. The base station includes a prioritymessage processor 320 shown in FIG. 91 and in FIG. 94, that receivesfrom the link control channel, the first message segment of the callcontrol message. The priority message processor 320 at the base stationthen resets a message segment buffer 322 in the base station and storesthe first segment of the call control message in the buffer 322, if thepriority interrupt flag “P” has a first value of one. The first value ofone for the priority interrupt flag “P” corresponds to a time criticalmessage segment. This operation effectively substitutes the more timecritical call control message for the first, station message at the basestation Z.

When the remaining segments of the call control message are received bythe base station's priority message processor 320 over the link controlchannel, the priority message processor 320 at the base stationconcatenates the remaining message segments with the first receivedmessage segment in the priority message buffer 322, since the priorityinterrupt flag “P” for those segments has a second value of zero. Thesecond value of zero for the priority interrupt flag “P” corresponds toa message segment that is not the first message segment in a messagehaving plural segments.

In an alternate embodiment of the invention, the base station's prioritymessage processor 320 of FIG. 94, selectively reassigns its messageprocessing capacity from low priority messages it is currentlytransmitting, to more time critical messages that it receives on thelink control channel. In accordance with the invention, the base stationZ is currently transmitting a transmitted spread signal comprising anoutgoing data traffic signal spread over a plurality of discrete trafficfrequencies and an outgoing message segment signal spread over aplurality of link control frequencies. This takes place during thetransmit interval of a time division duplex session with the remotestation X. The outgoing message segment signal from the base station ispart of a low priority message, such as a software download to theremote station. During the next receive interval of the time divisionduplex session, the base station Z receives a spread signal comprisingan incoming data traffic signal spread over a plurality of discretetraffic frequencies and an incoming message segment signal spread over aplurality of link control frequencies. The base station adaptivelydespreads the signals received at the base station by using despreadingweights in its spectral and spatial despreading processor 312 of FIG.91. Then the base station's priority message processor 320 of FIG. 94,detects a priority interrupt flag value “P” in the message segmentsignal.

In accordance with the alternate embodiment of the invention, the basestation's priority message processor 320 acts to reassign the station'smessage processing capacity by interrupting the next scheduledtransmission of the second outgoing message segment signal. An examplewhere this operation is required is when the remote station X sends acall control message to the base station that requires the base stationto quickly respond with a reply message. The priority message processor320 detects the priority interrupt flag P=1 in the first segment of thecall control message, and in response, resets the message segment buffer322 in the base station. The priority message processor 320 stores theincoming message segment signal in the message segment buffer 322. Thepriority message processor 320 of FIG. 94, determines that the callcontrol message received from the remote station requires a quick reply.In response to this, the priority message processor 320 at the basestation stores the last segment number sent for the low priorityoutgoing message it has been sending to the remote station. The basestation can then transmit its reply message to the remote station. Inthis manner, quick reply message can be sent in response to requestmessages. After the reply message has been sent by the base station, thelast segment number sent for the low priority outgoing message isretrieved by the priority message processor 320 and transmission isresumed by the base station, starting with the next segment number forthe low priority outgoing message. Alternately, the base stationconcatenates the incoming message segment signal with a previouslyreceived message segment, if the priority interrupt flag has a secondvalue of zero. The second value of zero for the priority interrupt flagcorresponds to a message segment that is not a first message segment ina message having plural segments. In this manner, the invention managesthe exchange of system management messages over the link control channelbetween a remote station and the base station so that time criticalsystem management messages are given priority over those that are nottime critical.

In FIG. 90, Alice and Bob each input data to remote station X. Thesender's traffic data is sent to the vector formation buffer 202 and thesender's system management information is sent to the priority messageprocessor 204, shown in greater detail in FIG. 92. Data vectors areoutput from buffer 202 to the trellis encoder 206. The data vectors arein the form of a 48-bit data message segment per transmit burst. The LCCvectors output from the priority message processor 204 to the trellisencoder 206 are in the form of a 48-bit priority message segment pertransmit burst, formed by concatenating a 47-bit message segment withthe one-bit priority interrupt flag. The trellis encoded data vectorsand LCC vectors are then output to the spectral spreading processor 208.The resultant data tones and LCC tones are then output from processor208 to the transmitter 210 for transmission to the base station.

The first four steps in the flow diagram 700 of FIG. 93 show the stepsat remote station X when it is the sender. The steps in the method oftransmission from a remote station to a base station are first for theRemote Station in step 710 to generate a priority message segment in thepriority message processor 204 of FIG. 92 and input it as a vector tothe link control channel. Then in step 720, the Remote Station performstrellis encoding of the link control channel vector and the datavectors. Then in Step 730, the Remote Station performs spectralspreading of the trellis encoded link control channel vector and datavectors. Then in Step 740, the Remote Station transmits the link controlchannel tone and data tones to the base station.

The personal wireless access network (PWAN) system described in thereferenced Alamouti, Stolarz, et al. patent application provides a moredetailed description of a high capacity mode, where one trafficpartition is used in one traffic channel. The Base transmits informationto multiple Remote Units in its cell. The transmission formats are for a64 kbits/sec traffic channel, together with a 4 kbps Link ControlChannel (LCC) between the Base and a Remote Unit. The binary sourcedelivers data to the sender's transmitter at 64 kbits/sec. Thistranslates to 48 bits in one transmission burst. The information bitsare encrypted according to a triple data encryption standard (DES)algorithm. The encrypted bits are then randomized in a datarandomization block. A bit to octal conversion block converts therandomized binary sequence into a sequence of 3-bit symbols. The symbolsequence is converted into 16 symbol vectors. The term vector generallyrefers to a column vector which is generally complex. One symbol fromthe LCC is added to form a vector of 17 symbols.

The 17-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the LCCsymbol). This process employs convolutional encoding that converts theinput symbol (an integer between 0 and 7) to another symbol (between 0and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is signal withinthe set of 16QAM (or 16PSK) constellation signals. (The term signal willgenerally refer to a signal constellation point.)

A link maintenance pilot signal (LMP) is added to form an 18-signalvector, with the LMP as the first elements of the vector. The resulting(18×1) vector is pre-multiplied by a (18×18) forward smearing matrix toyield a (18×1) vector b.

Vector b is element-wise multiplied by the (18×1) gain preemphasisvector to yield another (18×1) vector, c, where p denotes the trafficchannel index and is an integer. Vector c is post-multiplied by a (1×32)forward spatial and spectral spreading vector to yield a (18×32) matrixR(p). The number 32 results from multiplying the spectral spreadingfactor 4 and spatial spreading factor 8. The 18×32 matricescorresponding to all traffic channels carried (on the same trafficpartition) are then combined (added) to produce the resulting 18×32matrix S.

The matrix S is partitioned (by groups of four columns) into eight(18×4) submatrices (A₀ to A₇). (The indices 0 to 7, corresponds to theantenna elements over which these symbols will eventually betransmitted.) Each submatrix is mapped to tones within one trafficpartition.

A lower physical layer places the baseband signals in discrete Fouriertransfer (DFT) frequency bins where the data is converted into the timedomain and sent to its corresponding antenna elements (0 to 7) fortransmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next forward transmission burst.

FIG. 91 is an architectural diagram of the base station Z as a receiver.The data tones and LCC tones are received at the base station antennasA, B, C, and D. The receiver 310 passes the data tones and the LCC tonesto the spectral and spatial despreading processor 312. The despreadsignals are then output from the processor 312 to the trellis decoder314. The data vectors are then output to the vector disassembly buffer316. The LCC vectors are output to the priority message processor 320,shown in greater detail in FIG. 94. Alice's data and Bob's data areoutput from the buffer 316 to the public switched telephone network(PSTN). Priority message segments are passed from the priority messageprocessor 320 to the priority message buffer 322. There the segments areconcatenated into a full message 325 to be output on line 330.

The last five steps in the flow diagram 700 of FIG. 93, show the basestation Z as the receiver. In Step 750, the Base Station performsspectral and spatial despreading of the link control channel tone anddata tones. Then, in Step 760, the Base Station performs trellisdecoding of despread link control channel tone and data tones. Then inStep 770, the Base Station's priority message processor 320 determinesif the priority interrupt flag P=1. If it does, then priority messageprocessor 320 resets the priority message buffer 322 and loads the newlyreceived message segment in buffer 322. In step 780, alternately, if theBase Station's priority message processor 320 determines that thepriority interrupt flag P=0, then it concatenates the newly receivedmessage segment with the previously received message segment for thesame message in buffer 322. Then in Step 790, the priority messagebuffer 322 combines the plural message segments into a complete messageand outputs it on line 330. The completed message can be processedwithin the base station or it can be forwarded along with the receiveddata to the public switched telephone network.

In this manner, the invention manages the exchange of system managementmessages over the link control channel between a remote station and thebase station so that time critical system management messages are givenpriority over those that are not time critical.

Although the preferred embodiments of the invention have been describedin detail above, it will be apparent to those of ordinary skill in theart that obvious modifications may be made to the invention withoutdeparting from its spirit or essence. Consequently, the precedingdescription should be taken as illustrative and not restrictive, and thescope of the invention should be determined in view of the followingclaims.

TITLE OF THE INVENTION: “METHOD OF POLLING REMOTE STATIONS FORFUNCTIONAL QUALITY AND MAINTENANCE DATA IN A DISCRETE MULTITONE SPREADSPECTRUM COMMUNICATIONS SYSTEM”2455/4348 BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention involves improvements to communications systems andmethods in a wireless discrete multitone spread spectrum communicationssystem.

2. Description of Related Art

Wireless communications systems, such as cellular and personalcommunications systems, operate over limited spectral bandwidths. Theymust make highly efficient use of the scarce bandwidth resource toprovide good service to a large population of users. Code DivisionMultiple Access (CDMA) protocol has been used by wireless communicationssystems to efficiently make use of limited bandwidths. The protocol usesa unique code to distinguish each user's data signal from other users'data signals. Knowledge of the unique code with which any specificinformation is transmitted, permits the separation and reconstruction ofeach user's message at the receiving end of the communication channel.

Adaptive beamforming technology has become a promising technology forwireless service providers to offer large coverage, high capacity, andhigh quality service. Based on this technology, a wireless communicationsystem can improve its coverage capability, system capacity, andperformance significantly. The personal wireless access network (PWAN)system described in the referenced Alamouti, Stolarz, et al. patentapplications, uses adaptive beamforming combined with a form of the CDMAprotocol known as discrete multitone spread spectrum (DMT-SS) to provideefficient communications between a base station and a plurality ofremote units. Every effort must be made to avoid loading normal, highpriority traffic channels with system management information that has alower priority. An example of system management information is thefunctional quality and maintenance data for remote stations. Functionalquality data includes, for example, the history of thesignal-to-interference-and-noise ratio (SINR) or the history of the pathloss over a given measurement period for signals received at the remotestation from the base station. Maintenance data includes for example,self-test results and battery status at the remote station. Suchinformation is important for providing superior service to networkcustomers and for anticipating system failures. However, it is notnecessary to transmit such information when greater time criticalinformation such as call control messages need to be transmitted. Whatis needed is a way to communicate functional quality and maintenancedata from the remote stations to the base station without adverselyaffecting the transmission of messages having greater time criticality.

SUMMARY OF THE INVENTION

The invention disclosed herein is a new method to make the mostefficient use of the scarce spectral bandwidth in a wireless discretemultitone spread spectrum communications system. Each remote station inthe network collects functional quality and maintenance data for itself.During each data traffic session that a remote station has with the basestation, the remote station computes thesignal-to-interference-and-noise ratio (SINR) as a byproduct ofreceiving the discrete multitone spread spectrum signals from the basestation. The remote station stores the SINR data that it accumulates ina SINR history buffer. The remote station also computes the path loss ofthe signals received from the base station and stores the values itaccumulates in a path loss history buffer. The remote station runsself-test programs on a periodic basis and stores the results in aself-test results buffer. And the remote station monitors the status ofits backup battery and stores the status in a battery status buffer.Other functional quality and maintenance data can also be monitored andstored by the remote station.

In accordance with the invention, the base station periodicallytransmits a discrete multitone spread spectrum signal on the common linkchannel to each remote station, polling the respective remote station.The common link channel (CLC) is used by the base to transmit controlinformation to the remote stations. In response to the polling signal,the respective remote station activates its polling response processorto respond the poll. The polling response processor accesses the selftest buffer, the battery status buffer, the SINR history buffer, and thepath loss buffer to assemble a functional quality and maintenance datamessage. The functional quality and maintenance data message is thentransmitted back to the base station on the common access channel.

In accordance with the invention, the remote station prepares thefunctional quality and maintenance message for transmission over thecommon access channel of the network. The remote station forms a commonaccess channel vector that will be spread using the discrete multitonespread spectrum ( DMT-SS ) protocol to distribute the functional qualityand maintenance data message over a plurality of discrete tonefrequencies, forming a spread signal for the common access channel. TheCommon Access Channel (CAC) is used to transmit messages from the remotestation to the base. There is one grouping of tones assigned to eachchannel. These overhead channels are used in common by all of the remoteunits when they are exchanging control messages with the base station.

When the base station receives the functional quality and maintenancemessage on the common access channel tone from the remote station it haspolled, it performs spectral and spatial despreading of the signal andtrellis decoding of the signal to obtain a common access channel vectorbearing the functional quality and maintenance data. The functionalquality and maintenance message information is then stored in thefunctional quality and maintenance archive buffer, organized by eachresponding remote station.

A functional quality processor in the base station processes thefunctional quality and maintenance data to detect evidence ofsubstandard functioning of any channels between a remote processor andthe base station. If substandard functioning is detected, such as a lowSINR, then the functional quality processor updates the spreading anddespreading weights for the channels to improve the functional qualityof the traffic channels.

A maintenance processor in the base station processes the functionalquality and maintenance data to detect evidence of failing components ina remote processor. If any failing component is detected, such as a lowbattery, then the maintenance processor outputs a maintenance notice tothe system administrator.

The functional quality and maintenance data can initiate an alarm to beused for other realtime control. Or, the functional quality andmaintenance data can be logged for the compilation of a longer termreport of the traffic channel quality or the remote station's health.

In this manner, functional quality and maintenance data can becommunicated from the remote stations to the base station withoutadversely affecting the transmission of messages having greater timecriticality.

Currently, the invention has advantageous applications in the field ofwireless communications, such as cellular communications or personalcommunications, where bandwidth is scarce compared to the number of theusers and their needs. Such applications may be effected in mobile,fixed, or minimally mobile systems. However, the invention may beadvantageously applied to other, non-wireless, communications systems aswell.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 95 is an architectural diagram of the PWAN system, showing the basestation polling a remote station over the common link channel.

FIG. 96 is an architectural diagram of the PWAN system, showing theremote station transmitting a functional quality and maintenance messageto the base station over the common access channel.

FIG. 97 is an architectural diagram of the remote station X as a senderof functional quality and maintenance data.

FIG. 98 is an architectural diagram of the base station Z as a receiverof functional quality and maintenance data.

FIG. 99 is a flow diagram of the sequence of operational steps for theinvention.

DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 95 is an architectural diagram of the PWAN system, showing the basestation polling a remote station over the common link channel. FIG. 96shows the remote station transmitting a functional quality andmaintenance message to the base station over the common access channel.These are diagrams of the personal wireless access network (PWAN) systemdescribed in the referenced Alamouti, Stolarz, et al. patentapplications. Two users, Alice and a Bob, are located at the remotestation X and wish to transmit their respective data messages to thebase station Z. Station X is positioned to be equidistant from theantenna elements A, B, C, and D of the base station Z. Two other users,Chuck and Dave, are located at the remote station Y and also wish totransmit their respective data messages to the base station Z. Station Yis geographically remote from Station X and is not equidistant from theantenna elements A, B, C, and D of the base station Z. The remotestations X and Y and the base station Z use the form of the CDMAprotocol known as discrete multitone spread spectrum (DMT-SS) to provideefficient communications between the base station and the plurality ofremote station units. This protocol is designated in FIG. 95 asmulti-tone CDMA. In this protocol, the user's data signal is modulatedby a set of weighted discrete frequencies or tones. The weights arespreading weights that distribute the data signal over many discretetones covering a broad range of frequencies. The weights are complexnumbers with the real component acting to modulate the amplitude of atone while the complex component of the weight acts to modulate thephase of the same tone. Each tone in the weighted tone set bears thesame data signal. Plural users at the transmitting station can use thesame tone set to transmit their data, but each of the users sharing thetone set has a different set of spreading weights. The weighted tone setfor a particular user is transmitted to the receiving station where itis processed with despreading weights related to the user's spreadingweights, to recover the user's data signal. For each of the spatiallyseparated antennas at the receiver, the received multitone signals aretransformed from time domain signals to frequency domain signals.Despreading weights are assigned to each frequency component of thesignals received by each antenna element. The values of the despreadingweights are combined with the received signals to obtain an optimizedapproximation of individual transmitted signals characterized by aparticular multitone set and transmitting location.

The PWAN system has a total of 2560 discrete tones (carriers) equallyspaced in 8 MHZ of available bandwidth in the range of 1850 to 1990 MHZ.The spacing between the tones is 3.125 kHz. The total set of tones arenumbered consecutively form 0 to 2559 starting from the lowest frequencytone. The tones are used to carry traffic messages and overhead messagesbetween the base station and the plurality of remote units. The traffictones are divided into 32 traffic partitions, with each traffic channelrequiring at least one traffic partition of 72 tones.

In addition, the PWAN system uses overhead tones to establishsynchronization and to pass control information between the base stationand the remote units. A Common Link Channel (CLC) is used by the base totransmit control information to the Remote Units. A Common AccessChannel (CAC) is used to transmit messages from the Remote Unit to theBase. There is one grouping of tones assigned to each channel. Theseoverhead channels are used in common by all of the remote units whenthey are exchanging control messages with the base station.

In the PWAN system, Time Division Duplexing (TDD) is used by the basestation and the remote unit to transmit data and control information inboth directions over the same multi-tone frequency channel. Transmissionfrom the base station to the remote unit is called forward transmissionand transmission from the remote unit to the base station is calledreverse transmission. The time between recurrent transmissions fromeither the remote unit or the base station is the TDD period. In everyTDD period, there are four consecutive transmission bursts in eachdirection. Data is transmitted in each burst using multiple tones. Thebase station and each remote unit must synchronize and conform to theTDD timing structure and both the base station and the remote unit mustsynchronize to a framing structure. All remote units and base stationsmust be synchronized so that all remote units transmit at the same timeand then all base stations transmit at the same time. When a remote unitinitially powers up, it acquires synchronization from the base stationso that it can exchange control and traffic messages within theprescribed TDD time format. The remote unit must also acquire frequencyand phase synchronization for the DMT-SS signals so that the remote isoperating at the same frequency and phase as the base station.

Selected tones within each tone set are designated as pilots distributedthroughout the frequency band. Pilot tones carry known data patternsthat enable an accurate channel estimation. The series of pilot tones,having known amplitudes and phases, have a known level and are spacedapart by approximately 30 KHz to provide an accurate representation ofthe channel response (i.e., the amplitude and phase distortionintroduced by the communication channel characteristics) over the entiretransmission band.

In accordance with the invention, a new method makes the most efficientuse of the scarce spectral bandwidth in a wireless discrete multitonespread spectrum communications system. Each remote station in thenetwork collects functional quality and maintenance data for itself.During each data traffic session that a remote station has with the basestation, the remote station X of FIG. 97 computes thesignal-to-interference-and-noise ratio (SINR) as a byproduct ofreceiving the discrete multitone spread spectrum signals from the basestation Z. The remote station stores the SINR data that it accumulatesin a SINR history buffer 224. The remote station also computes the pathloss of the signals received from the base station and stores the valuesit accumulates in a path loss history buffer 226. The remote stationruns self-test programs on a periodic basis and stores the results in aself-test results buffer 220. And the remote station monitors the statusof its backup battery and stores the status in a battery status buffer222. Other functional quality and maintenance data can also be monitoredby the remote station and stored in buffers.

In accordance with the invention, the base station Z periodicallytransmits a discrete multitone spread spectrum (DMT-SS) signal on thecommon link channel to each remote station, polling the respectiveremote station, as shown in FIG. 95. The common link channel (CLC) isused by the base to transmit control information to the remote stations.Simultaneously, data traffic from the public switched telephone network(PSTN) arrives at the base station Z and is converted into data trafficDMT-SS tones which are transmitted to the remote stations. In responseto the base station's polling signal being received by the remotestation X at its input 230, the respective remote station of FIG. 97,activates its polling response processor 228 to respond the poll. Thepolling response processor 228 accesses the self test buffer 220, thebattery status buffer 222, the SINR history buffer 224, and the pathloss buffer 226 to assemble a functional quality and maintenance datamessage. The message is formed into a common access channel vector thatis input to the trellis encoder 206 and then to the spectral spreadingprocessor 208 to produce the common access channel tone. The commonaccess channel tone with the functional quality and maintenance datamessage is then transmitted by transmitter 210 as a DMT-SS signal backto the base station Z on the common access channel.

In accordance with the invention, the remote station X prepares thefunctional quality and maintenance message for transmission over thecommon access channel of the network. The remote station forms a commonaccess channel vector that will be spread using the discrete multitonespread spectrum (DMT-SS) protocol to distribute the functional qualityand maintenance data message over a plurality of discrete tonefrequencies, forming a spread signal for the common access channel. Thecommon access channel (CAC) is used to transmit messages from the remotestation to the base. There is one grouping of tones assigned to eachchannel. These overhead channels are used in common by all of the remoteunits when they are exchanging control messages with the base station.

When the base station Z of FIG. 98 receives the functional quality andmaintenance message on the common access channel tone from the remotestation X that it has polled, it performs spectral and spatialdespreading of the signal in the spectral and spatial despreadingprocessor 312 and trellis decoding of the signal in the trellis decoder314 to obtain a common access channel vector bearing the functionalquality and maintenance data. The functional quality and maintenancedata are then stored in the functional quality and maintenance archivebuffer 320, organized by each responding remote station.

A functional quality processor 322 in the base station Z processes thefunctional quality data to detect evidence of substandard functioning ofany channels between a remote station and the base station. Ifsubstandard functioning is detected, such as a low SINR, then thefunctional quality processor 322 updates the spreading and despreadingweights in buffer 340 for the channels to improve the functional qualityof the channels, either traffic or overhead channels.

A maintenance processor 330 in the base station Z processes themaintenance data to detect evidence of failing components in a remotestation. If any failing component is detected, such as a low battery,then the maintenance processor 330 outputs a maintenance notice data 350to the system administrator.

The functional quality and maintenance data can initiate an alarm to beused for other realtime control. Or, the functional quality andmaintenance data can be logged for the compilation of a longer termreport of the traffic channel quality or the remote station's health.

In this manner, functional quality and maintenance data can becommunicated from the remote stations to the base station withoutadversely affecting the transmission of messages having greater timecriticality.

The personal wireless access network (PWAN) system described in thereferenced Alamouti, Stolarz, et al. patent applications provides a moredetailed description of a high capacity mode, where one trafficpartition is used in one traffic channel. The Base transmits informationto multiple Remote Units in its cell. The transmission formats are for a64 kbits/sec traffic channel, together with a 4 kbps link controlchannel (LCC) between the base and a remote station. The binary sourcedelivers data to the sender's transmitter at 64 kbits/sec. Thistranslates to 48 bits in one transmission burst. The information bitsare encrypted according to a triple data encryption standard (DES)algorithm. The encrypted bits are then randomized in a datarandomization block. A bit to octal conversion block converts therandomized binary sequence into a sequence of 3-bit symbols. The symbolsequence is converted into 16 symbol vectors. The term vector generallyrefers to a column vector which is generally complex. One symbol fromthe LCC is added to form a vector of 17 symbols.

The 17-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the LCCsymbol). This process employs convolutional encoding that converts theinput symbol (an integer between 0 and 7) to another symbol (between 0and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is signal withinthe set of 16QAM (or 16PSK) constellation signals. (The term signal willgenerally refer to a signal constellation point.)

A link maintenance pilot signal (LMP) is added to form an 18-signalvector, with the LMP as the first elements of the vector. The resulting(18×1) vector is pre-multiplied by a (18×18) forward smearing matrix toyield a (18×1) vector b.

Vector b is element-wise multiplied by the (18×1) gain preemphasisvector to yield another (18×1) vector, c, where p denotes the trafficchannel index and is an integer. Vector c is post-multiplied by a (1×32)forward spatial and spectral spreading vector to yield a (18×32) matrixR(p). The number 32 results from multiplying the spectral spreadingfactor 4 and spatial spreading factor 8. The 18×32 matricescorresponding to all traffic channels carried (on the same trafficpartition) are then combined (added) to produce the resulting 18×32matrix S.

The matrix S is partitioned (by groups of four columns) into eight(18×4) submatrices (A₀ to A₇). (The indices 0 to 7, corresponds to theantenna elements over which these symbols will eventually betransmitted.) Each submatrix is mapped to tones within one trafficpartition.

A lower physical layer places the baseband signals in discrete Fouriertransfer (DFT) frequency bins where the data is converted into the timedomain and sent to its corresponding antenna elements (0 to 7) fortransmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next forward transmission burst.

FIG. 99 is a flow diagram 700 of the sequence of operational steps forthe invention. In step 710, the remote station monitors and buffers thefunctional quality data, including the SINR and path loss for sessionswith the base station. In step 720, the remote station monitors andbuffers the maintenance data, including self-test results and batterystatus, for the remote station. In step 730, the base station transmitsa polling signal on the common link channel tone to the remote station.In step 740, the remote station accesses the functional quality data andthe maintenance data from its buffers, assembles the data into a messagevector, and transmits it on the common access channel tone to the basestation. The remote station simultaneously transmits data trafficchannel tones to the base station. In step 750, the base stationperforms spectral and spatial despreading of the common access channeltone and the data traffic tones. In step 760, the base station performstrellis decoding to recover the common access channel vector bearing thefunctional quality and maintenance message. In step 770, the basestation archives the functional quality and maintenance data. In step780, the base station analyzes the functional quality data and updatesthe despreading and spreading weights to maximize the quality of thechannels it establishes with the remote station. In step 790, the basestation analyzes the maintenance data and outputs maintenance notices torepair or replace failing components at the remote station. In thismanner, functional quality and maintenance data can be communicated fromthe remote stations to the base station without adversely affecting thetransmission of messages having greater time criticality.

Although the preferred embodiments of the invention have been describedin detail above, it will be apparent to those of ordinary skill in theart that obvious modifications may be made to the invention withoutdeparting from its spirit or essence. Consequently, the precedingdescription should be taken as illustrative and not restrictive, and thescope of the invention should be determined in view of the followingclaims.

TITLE OF THE INVENTION: “POWER MANAGEMENT METHOD FOR A DISCRETEMULTITONE SPREAD SPECTRUM COMMUNICATIONS SYSTEM”2455/4382 BACKGROUND OFTHE INVENTION

1. Field of the Invention

This invention involves improvements to communications systems andmethods in a wireless discrete multitone spread spectrum communicationssystem.

2. Description of Related Art

Wireless communications systems, such as cellular and personalcommunications systems, operate over limited spectral bandwidths. Theymust make highly efficient use of the scarce bandwidth resource toprovide good service to a large population of users. Code DivisionMultiple Access (CDMA) protocol has been used by wireless communicationssystems to efficiently make use of limited bandwidths. The protocol usesa unique code to distinguish each user's data signal from other users'data signals. Knowledge of the unique code with which any specificinformation is transmitted, permits the separation and reconstruction ofeach user's message at the receiving end of the communication channel.

The personal wireless access network (PWAN) system described in thereferenced Alamouti, Stolarz, et al. patent application, uses a form ofthe CDMA protocol known as discrete multitone spread spectrum ( DMT-SS )to provide efficient communications between a base station and aplurality of remote units. In this protocol, the user's data signal ismodulated by a set of weighted discrete frequencies or tones. Theweights are spreading codes that distribute the data signal over manydiscrete tones covering a broad range of frequencies. The weights arecomplex numbers with the real component acting to modulate the amplitudeof a tone while the complex component of the weight acts to modulate thephase of the same tone. Each tone in the weighted tone set bears thesame data signal. Plural users at the transmitting station can use thesame tone set to transmit their data, but each of the users sharing thetone set has a different set of spreading codes. The weighted tone setfor a particular user is transmitted to the receiving station where itis processed with despreading codes related to the user's spreadingcodes, to recover the user's data signal. For each of the spatiallyseparated antennas at the receiver, the received multitone signals aretransformed from time domain signals to frequency domain signals.Despreading weights are assigned to each frequency component of thesignals received by each antenna element. The values of the despreadingweights are combined with the received signals to obtain an optimizedapproximation of individual transmitted signals characterized by aparticular multitone set and transmitting location. The PWAN system hasa total of 2560 discrete tones (carriers) equally spaced in 8 MHZ ofavailable bandwidth in the range of 1850 to 1990 MHZ. The spacingbetween the tones is 3.125 kHz. The total set of tones are numberedconsecutively form 0 to 2559 starting from the lowest frequency tone.The tones are used to carry traffic messages and overhead messagesbetween the base station and the plurality of remote units. The traffictones are divided into 32 traffic partitions, with each traffic channelrequiring at least one traffic partition of 72 tones.

In addition, the PWAN system uses overhead tones to establishsynchronization and to pass control information between the base stationand the remote units. A Common Link Channel (CLC) is used by the base totransmit control information to the Remote Units. A Common AccessChannel (CAC) is used to transmit messages from the Remote Unit to theBase. There is one grouping of tones assigned to each channel. Theseoverhead channels are used in common by all of the remote units whenthey are exchanging control messages with the base station.

In the PWAN system, Time Division Duplexing (TDD) is used by the basestation and the remote unit to transmit data and control information inboth directions over the same multi-tone frequency channel. Transmissionfrom the base station to the remote unit is called forward transmissionand transmission from the remote unit to the base station is calledreverse transmission. The time between recurrent transmissions fromeither the remote unit or the base station is the TDD period. In everyTDD period, there are four consecutive transmission bursts in eachdirection. Data is transmitted in each burst using multiple tones. Thebase station and each remote unit must synchronize and conform to theTDD timing structure and both the base station and the remote unit mustsynchronize to a framing structure. All remote units and base stationsmust be synchronized so that all remote units transmit at the same timeand then all base stations transmit at the same time. When a remote unitinitially powers up, it acquires synchronization from the base stationso that it can exchange control and traffic messages within theprescribed TDD time format. The remote unit must also acquire frequencyand phase synchronization for the DMT-SS signals so that the remote isoperating at the same frequency and phase as the base station.

In the PWAN system, some of the tone frequencies are pilot tones used totransmit known symbols from the base station to the remote or from theremote station to the base, to enable synchronization of the stations.The referenced Hoole and the referenced Veintimilla patent applicationsdiscuss some of these features.

The PWAN system performs matrix operations involved in null-steering andcode-nulling procedures. The retrodirective weights calculated from thedespreading weights by the PWAN system are provided to transmitter pathfor use during data spreading, beam forming, and generating inverse fastFourier transforms (IFFTs).

In accordance with one aspect of the PWAN system, adaptive antennaarrays are used in conjunction with a beam forming algorithm to achievespatial diversity within each cell and implement SDMA. That is, signalsoutput by the antennas are directionally formed by selectivelyenergizing different antenna sensors with different signal gains so thatremote terminals in one portion of a cell are able to communicate withthe base station while other remote terminals in a different portion ofthe cell may communicate with the same base station, despite the factthat they are using the same tone set and code. It should be understoodthat in the fixed implementation of the current PWAN system, i.e., wherethe remote access terminals do not move substantially duringcommunication with the base station, usually staying within a cellduring communication, the beam forming algorithm used in the airlinkneed not account for mobile remote units leaving and entering the cell.In one advantageous embodiment, each cell is partitioned into foursectors where each sector transmits and receives over one of the foursub-band pairs.

As set forth above, the beam forming method of the PWAN system, like theuse of codes, should not be conceived as separate from the overalladaptive equalization method of the PWAN system. Rather, the method usedto selectively energize the antenna sensors (during transmission) orselectively weight the signals received on the different sensor elements(during reception) is subsumed into the overall method used to maximizeSINR. The relation of the beam forming method to the overallmaximization of SIR method will be described in greater detail below.

The use of spread-spectrum technology (particularly DMT-SS) anddirectional antennas within the preferred airlink of the PWAN systemallows for several error cancellation benefits, including effects thatare analogous to code nulling and null steering, by means of linearweighting in code and space.

Code-nulling is used to discriminate between non-orthogonal signalsemanating from adjacent cells. Again, the code-nulling method should beunderstood in the context of the maximization of SINR method of the PWANsystem. That is, the code-nulling method should be considered as theportion of the method that maximizes SINR with respect to the codedomain.

It should be understood that if signals generated within the same cellor beam all have orthogonal spreading codes, code-nulling is typicallynot necessary since the orthogonality is sufficient to ensure that thereis no cross modulation. However, as mentioned above, the spreading codesused within a particular cell may not be orthogonal, although they arepreferably linearly independent. Furthermore, the transceivers withinthe neighboring cells may employ spreading codes that have a randomcorrelation with the spreading codes used in the local cell.

By adjusting the spreading weights associated with each communicationschannel the base station is able to cross-correlate these signals on thesame tone set to subtract out interference due to “neighboring” signals.In one embodiment, the base station has the spreading codes used tospread different signals assigned to the same tone set, so that thisinformation can be used to initially calculate the appropriate weightsfor nulling out interference from other codes.

As discussed above, when the spreading codes used to spread distinctdata signals are orthogonal, the spread data can be precisely recoveredduring despreading. However, when the spreading codes are not orthogonal(as is the case with spreading codes that are used in neighboringcells), cross modulation may result so that the data signals are notable to be precisely distinguished by simple despreading (i.e.,despreading without code-nulling).

In order to compensate for this phenomenon, code-nulling weights aremultiplied by the received signal vector. By nulling out the crossmodulation present in the received signal, the appropriate values of thedata bits are output by the receiver. As long as the complex spreadingweights are linearly independent the exact symbol values can bediscriminated by this method. It will be appreciated that thecode-nulling procedure above is inherently implemented during derivationof the overall weights that maximize the SINR.

In addition to code-nulling, the directional antenna forms signalsincluding null regions (i.e., regions where the antenna attenuatesincoming signals or where there is a very low antenna gain). These nullregions can be formed in a pattern so that the nulls are directedtowards known interferers (e.g., interfering signal sources orinterfering multi-path reflectors). In this manner, interfering signalsare de-emphasized in the spatial domain. As will be discussed in greaterdetail below, the use of null-steering in conjunction with code-nullingis highly advantageous.

In accordance with one aspect of the PWAN system, significant processingtime and sophistication can be saved since significant similarity existsbetween the methods for performing null-steering and code-nulling.Specifically, the mathematical formalism used to achieve null-steeringis analogous to the formalism used to achieve code-nulling. According tothis analogy, just as the tones in a tone set are multiplied by complexweights to alter the amplitude and phase of the tones, so are the gainand relative phase of signals output and received by the antennaelements altered by a set of multiplicative weights. This multiplicationby complex weights can be expressed in a matrix form for both codenulling—a spectral concept—and null steering—a spatial concept. Thus,the calculations performed in the spectral code domain correspondformally to the calculations performed in the spatial domain.Consequently, null steering can be performed in a system usingcode-nulling simply by adding an extra dimension to the matrices usedfor calculating the complex weights and multiplying the signals by theseweights.

What is needed in wireless communications systems composed of multiplecells, is the ability to control the power level of signals transmittedby remote stations and base stations minimize interference whileinsuring that signals reach their intended destination.

SUMMARY OF THE INVENTION

The invention enables control over the power level of signalstransmitted by remote stations and base stations in a DMT-SS wirelessnetwork, to minimize interference while insuring that signals reachtheir intended destination. In accordance with the invention, the basestation begins by transmitting forward pilot tones with a prearrangedinitial forward signal power level, to the remote station. The signalreceived by the remote station has a signal power level that is lessthan the prearranged initial forward signal power level, the differencebeing a measure of the channel loss between the base station and theremote station. The remote station stores the value of the channel lossit measures. Then the remote station continues by transmitting reversepilot tones with a prearranged initial reverse signal power level, tothe base station. The signal received by the base station has a signalpower level that is less than the prearranged initial reverse signalpower level, the difference being a measure of the channel loss betweenthe base station and the remote station. The base station stores thevalue of the channel loss it measures.

The base station prepares despreading weights to despread the DMT-SSsignals it receives from the remote station. Then the base uses theprinciple of retrodirectivity to compute spreading weights fortransmission of DMT-SS signals to the remote station. The spreadingweights calculated at the base station include a factor based on themeasured channel loss stored at the base station, to overcome thechannel loss so that forward signals transmitted to the remote stationwill arrive there with a desired received signal power level.

The remote station prepares despreading weights to despread the DMT-SSsignals it receives from the base station. Then the remote station usesthe principle of retrodirectivity to compute spreading weights fortransmission of DMT-SS signals to the base station. The spreadingweights calculated at the remote station include a factor based on themeasured channel loss stored at the remote station, to overcome thechannel loss so that reverse signals transmitted to the base stationwill arrive there with a desired received signal power level.

In this manner, the invention controls the power level of signalstransmitted by remote stations and base stations to minimizeinterference while insuring that signals reach their intendeddestination.

Currently, the invention has advantageous applications in the field ofwireless communications such as cellular communications or personalcommunications, where bandwidth scarce compared to the number of theusers and their needs. Such applications may be effected in mobile,fixed or minimally mobile systems. However, the invention may beadvantageously applied to other, non-wireless, communications systems aswell.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 100 is an architectural diagram of the personal wireless accessnetwork (PWAN), showing the base station Z transmitting forward pilottones with a prearranged initial forward signal power level, to theremote station X and to the remote station Y.

FIG. 101 is an architectural diagram of the personal wireless accessnetwork (PWAN) of FIG. 100, showing the remote station X transmittingreverse pilot tones with a prearranged initial reverse signal powerlevel, to the base station Z.

DISCUSSION OF THE PREFERRED EMBODIMENT

FIG. 100 is an architectural diagram of the personal wireless accessnetwork (PWAN), showing the base station Z transmitting forward pilottones with a prearranged initial forward signal power level, to theremote station X and to the remote station Y. The signal received by theremote station X has a signal power level that is less than theprearranged initial forward signal power level, the difference being ameasure of the channel loss between the base station and the remotestation X. The remote station stores the value of the channel loss itmeasures.

FIG. 101 is an architectural diagram of the personal wireless accessnetwork (PWAN) of FIG. 100, showing the remote station X transmittingreverse pilot tones with a prearranged initial reverse signal powerlevel, to the base station Z. The signal received by the base station Zhas a signal power level that is less than the prearranged initialreverse signal power level, the difference being a measure of thechannel loss between the base station and the remote station X. The basestation stores the value of the channel loss it measures. The basestation includes a retrodirective power management unit. The baseprepares despreading weights to despread the DMT-SS signals it receivesfrom the remote station X. Then the base uses the principle ofretrodirectivity to compute spreading weights for transmission of DMT-SSsignals to the remote station X. The spreading weights calculated at thebase station include a factor based on the measured channel loss storedat the base station, to overcome the channel loss so that forwardsignals transmitted to the remote station X will arrive there with adesired received signal power level.

The remote station includes a retrodirective power management unit. Theremote station prepares despreading weights to despread the DMT-SSsignals it receives from the base station Z. Then the remote station Xuses the principle of retrodirectivity to compute spreading weights fortransmission of DMT-SS signals to the base station Z. The spreadingweights calculated at the remote station X include a factor based on themeasured channel loss stored at the remote station X, to overcome thechannel loss so that reverse signals transmitted to the base station Zwill arrive there with a desired received signal power level.

The resulting invention enables control over the power level of signalstransmitted by remote stations and base stations to minimizeinterference while insuring that signals reach their intendeddestination.

FIG. 100 illustrates the personal wireless access network (PWAN) systemdescribed in the referenced Alamouti, et al. patent application. Twousers, Alice and Bob, are located at the remote station X and willexchange their respective data messages with the base station Z. StationX is positioned to be equidistant from the antenna elements A and B, ofthe base station Z. Two other users, Chuck and Dave, are located at theremote station Y and also will exchange their respective data messageswith the base station Z. Station Y is geographically remote from StationX and is not equidistant from the antenna elements A and B of the basestation Z. The remote stations X and Y and the base station Z use theform of the CDMA protocol known as discrete multitone spread spectrum(DMT-SS) to provide efficient communications between the base stationand the plurality of remote station units. This protocol is designatedin FIG. 100 as multi-tone CDMA. In this protocol, the user's data signalis modulated by a set of weighted discrete frequencies or tones. Theweights are spreading weights that distribute the data signal over manydiscrete tones covering a broad range of frequencies. The weights arecomplex numbers with the real component acting to modulate the amplitudeof a tone while the complex component of the weight acts to modulate thephase of the same tone. Each tone in the weighted tone set bears thesame data signal. Plural users at the transmitting station can use thesame tone set to transmit their data, but each of the users sharing thetone set has a different set of spreading weights. The weighted tone setfor a particular user is transmitted to the receiving station where itis processed with despreading weights related to the user's spreadingweights, to recover the user's data signal. For each of the spatiallyseparated antennas at the receiver, the received multitone signals aretransformed from time domain signals to frequency domain signals.Despreading weights are assigned to each frequency component of thesignals received by each antenna element. The values of the despreadingweights are combined with the received signals to obtain an optimizedapproximation of individual transmitted signals characterized by aparticular multitone set and transmitting location. The PWAN system hasa total of 2560 discrete tones (carriers) equally spaced in 8 MHz ofavailable bandwidth in the range of 1850 to 1990 MHz. The spacingbetween the tones is 3.125 kHz. The total set of tones are numberedconsecutively form 0 to 2559 starting from the lowest frequency tone.The tones are used to carry traffic messages and overhead messagesbetween the base station and the plurality of remote units. The traffictones are divided into 32 traffic partitions, with each traffic channelrequiring at least one traffic partition of 72 tones.

The PWAN system has a total of 2560 discrete tones (carriers) equallyspaced in 8 MHz of available bandwidth in the range of 1850 to 1990 MHz.The spacing between the tones is 3.125 kHz. The total set of tones arenumbered consecutively form 0 to 2559 starting from the lowest frequencytone. The tones are used to carry traffic messages and overhead messagesbetween the base station and the plurality of remote units. The traffictones are divided into 32 traffic partitions, with each traffic channelrequiring at least one traffic partition of 72 tones.

In addition, the PWAN system uses overhead tones to establishsynchronization and to pass control information between the base stationand the remote units. A Common Link Channel (CLC) is used by the base totransmit control information to the Remote Units. A Common AccessChannel (CAC) is used to transmit messages from the Remote Unit to theBase. There is one grouping of tones assigned to each channel. Theseoverhead channels are used in common by all of the remote units whenthey are exchanging control messages with the base station.

In the PWAN system, Time Division Duplexing (TDD) is used by the basestation and the remote unit to transmit data and control information inboth directions over the same multi-tone frequency channel. Transmissionfrom the base station to the remote unit is called forward transmissionand transmission from the remote unit to the base station is calledreverse transmission. The time between recurrent transmissions fromeither the remote unit or the base station is the TDD period. In everyTDD period, there are four consecutive transmission bursts in eachdirection. Data is transmitted in each burst using multiple tones. Thebase station and each remote unit must synchronize and conform to theTDD timing structure and both the base station and the remote unit mustsynchronize to a framing structure. All remote units and base stationsmust be synchronized so that all remote units transmit at the same timeand then all base stations transmit at the same time. When a remote unitinitially powers up, it acquires synchronization from the base stationso that it an exchange control and traffic messages within theprescribed TDD time format. The remote unit must also acquire frequencyand phase synchronization for the DMT-SS signals so that the remote isoperating at the same frequency and phase as the base station.

Selected tones within each tone set are designated as pilots distributedthroughout the frequency band. Pilot tones carry known data patternsthat enable an accurate channel estimation. The series of pilot tones,having known amplitudes and phases, have a known level and are spacedapart by approximately 30 kHz to provide an accurate representation ofthe channel response (i.e., the amplitude and phase distortionintroduced by the communication channel characteristics) over the entiretransmission band.

The personal wireless access network (PWAN) system described in thereferenced Alamouti, et al. patent application provides a more detaileddescription of the system. The Base transmits information to multipleRemote Units in its cell. The transmission formats are for a 64kbits/sec traffic channel, together with a 4 kbps Link Control Channel(LCC) between the Base and a Remote Unit. The binary source deliversdata to the sender's transmitter at 64 kbits/sec. This translates to 48bits in one transmission burst. The information bits are encryptedaccording to a triple data encryption standard (DES) algorithm. Theencrypted bits are then randomized in a data randomization block. A bitto octal conversion block converts the randomized binary sequence into asequence of 3-bit symbols. The symbol sequence is converted into 16symbol vectors. The term vector generally refers to a column vectorwhich is generally complex. One symbol from the LCC is added to form avector of 17 symbols.

The 17-symbol vector is trellis encoded. The trellis encoding startswith the most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the LCCsymbol). This process employs convolutional encoding that converts theinput symbol (an integer between 0 and 7) to another symbol (between 0and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is signal withinthe set of 16QAM (or 16PSK) constellation signals. (The term signal willgenerally refer to a signal constellation point.)

A link maintenance pilot signal (LMP) is added to form an 18-signalvector, with the LMP as the first elements of the vector. The resulting(18×1) vector is pre-multiplied by a (18×18) forward smearing matrix toyield a (18×1) vector b.

Vector b is element-wise multiplied by the (18×1) gain preemphasisvector to yield another (18×1) vector, c, where p denotes the trafficchannel index and is an integer. Vector c is post-multiplied by a (1×32)forward spatial and spectral spreading vector to yield a (18×32) matrixR(p). The number 32 results from multiplying the spectral spreadingfactor 4 and spatial spreading factor 8. The 18×32 matricescorresponding to all traffic channels carried (on the same trafficpartition) are then combined (added) to produce the resulting 18×32matrix S.

The matrix S is partitioned (by groups of four columns) into eight(18×4) submatrices (A₀ to A₇). (The indices 0 to 7, corresponds to theantenna elements over which these symbols will eventually betransmitted.) Each submatrix is mapped to tones within one trafficpartition.

A lower physical layer places the baseband signals in discrete Fouriertransfer (DFT) frequency bins where the data is converted into the timedomain and sent to its corresponding antenna elements (0 to 7) fortransmission over the air.

This process is repeated from the start for the next 48 bits of binarydata to be transmitted in the next forward transmission burst.

The reverse channel transmission from the remote stations is analogousto the forward channel transmission from the base station.

This section specifies the smearing matrix C_(rev−smear). The input tothe smearing block is the (18×1) vector D_(rev). The output of thesmearing operation (vector b) can then be described by the matrixmultiplication of d_(rev) and the (18×18) smearing matrix C_(rev−smear).That is

b=C _(rev=smear) d _(rev)

C_(fwd−smear) is the constant valued matrix shown below (see thereferenced Alamouti, et al application)

where,

α=(ρ_(LMP)/(1+ρ_(LMP)))^(½)

β=(1/(1+ρ_(LMP)))

ρ_(LMP) is the ratio of pilot to data power that is a physical layerprovisionable parameter whose value is nominally set to one.

The δ_(i)is are elements of the cluster scrambling vector δ_(smear) thatis unique to the Remote Unit. δ_(smear) is a 17-element vector that isused to ensure that the smeared data from one user received in aparticular traffic partition at the Base is uncorrelated with otherusers within the same traffic partition in the local cell and adjacentcells. δ_(smear) is given by(see the referenced Alamouti, et al.application).

The ith element of δ_(smear) has the form e^(jφ) _(smear) ^((i)) where_(100 smear)(i) is a real number between 0 and 2π generated by apseudo-random number generator creating unique sequences for each RemoteUnit. The details of the pseudo-random number generator areimplementation dependent and need not be known at the Base.

This section defines the (1×4) reverse spectral spreading vector g^(H)_(rev). The input to the spectral spreading block is the (18×1) vectorb. The output of the spectral and spatial spreading operation, (18×4)matrix S_(rev), is the matrix multiplication of b and the (1×4) Spectralspreading vector g^(H) _(rev):

S _(rev) =bg ^(H) _(rev)

where,

g ^(H) _(rev) =[g 0 g 1 g 2 . . . g 30 g 31]

The elements of vector g^(H) _(rev) are transmit spreading weightscalculated throughout the transmission. The algorithm for the derivationof these weights is implementation dependent. However, to clarify theprocedure a specific algorithm for the derivation of these weights isdescribed below.

The Remote Unit derives its new transmit weights based on the mostrecent data received on the forward channel. The transmit weights are ascaled version of the received weights using four receive frequenciesfor a single antenna.

The receive weight vector w^(H) _(fwd) has four elements (w₀−w₃) thatare mapped to spectral components.

For the Remote Unit traffic establishment procedure, the transmitweights (g₀−g₃) are calculated according to the following equation:

g ^(H) _(rev)(p)=α_(rev)(n)π_(rev) w ^(H) _(fwd)

where α_(fwd)(n) is the Base gain ramp-up factor for the nth packet andwhere B_(rev) is the Remote Unit power management factor defined by theequation below:

π _(rev)=λ_(p) K _(fwd)+(1−λ_(p))K _(rev)(π_(loss)(n,p)/abs(w_(fwd)(p)))

where,

λ_(p) is the exponential decay or “forget factor” nominally set to 0.97

π_(loss) is the reciprocal of the Base-Remote Unit channel gain measuredusing the Remote Unit Synchronization Pilot (RSP) tones

K_(rev) is the target Base receive power (nominally −103 dBm)

n is the burst index

p is the link index

For the Remote Unit traffic establishment procedure, the receive weightsare adaptively calculated using the following equation:

 w _(fwd) =R ⁻¹ _(xx) r _(xd)

where

w_(fwd) is the (4×1) receive weight vector

r_(xd) is an estimate of the (4×1) cross-correlation vector of thereceived (4×1) vector x and the LMP (or the desired data) d

R⁻¹ _(xx) is an estimate of the (4×4) inverted auto-correlation matrixof the received vector x

For the Remote Unit steady-state procedure, receive weights areadaptively calculated using the following equation:

w _(rev) =R ⁻¹ _(xx) r _(xy)

where

w_(fwd) is the (4×1) weight vector

r_(xy) is an estimate of the (4×1) cross-correlation vector of thereceived (4×1) vector x and the despread data y

R⁻¹ _(xx) is an estimate of the (4×4) inverted auto-correlation matrixof the received vector x

The receive weights (w₀−w₃) are mapped to spectral components. Thetransmit weights (g₀−g₃) are a scaled version of the receive weights.The scaling is made according to the following equation:

g ^(H) _(rev)(p)=B _(rev) w ^(H) _(fwd)

where π_(rev) is the Remote Unit power management factor definedearlier.

Correlation estimates are computed over four forward-channel burst. Thenew despreading weights are applied to four forward channel bursts withno delay. The spreading weights are applied to eight reverse channelbursts after an 8-burst delay. Correlation estimates are made using anexponentially average block summation. The exponential decay constant isprovisional with a nominal value of 0.7.

The power output characteristics of Base transmissions on the forwardchannel are different from that of the Remote Unit transmissions on thereverse channel. The forward channel transmission from a Base to a givenRemote Unit is maintained at a fixed power level during the duration ofa connection. The power level is determined by the Base RME prior to thestart of the connection using a power management algorithm.

A forward RF channel transmission is initiated by a 180 ms ramp-upperiod (240 forward channel bursts) during the traffic establishmentperiod. The ramp-up starts after a connection is established between theBase and a given Remote Unit. The data transmitted during this periodare known link maintenance pilots. The maximum (steady state) power isreached after 240 channel bursts (180 msec) and maintained throughoutthe connection.

The following equation shows the forward channel ramp-up schedulerelative to the steady state power,

α_(fwd)(n)=(1−e ^(−5(8[n/8]))(1−e ⁻⁵))²

for n<240

α_(fwd)(n)=1

otherwise

where n is the forward channel burst number relative to the start of thetransmission.

The reverse channel transmissions from a Remote Unit to its Base isadaptively varied to ensure that the received power from all RUs attheir Base is maintained at a relatively constant level. The Remote Unitpower management algorithm is implementation dependent. One example ofthe algorithm is discussed in the Section on the Reverse Channel Format.

A reverse RF channel transmission is initiated by a 180 ms ramp-upperiod (240 reverse channel bursts) during the traffic establishmentperiod. The ramp-up starts after a connection is established between theRemote Unit and its Base. The data transmitted during this period areknown LMPs. The maximum (steady state) power is reached after 240reverse channel bursts (180 msec).

The following equation shows the reverse channel ramp-up schedulerelative to the steady state power,

α_(rev)(n)=(1−e ^(−5(8[n/8]))/(1−e ⁻⁵))²

for n<240

α_(rev)(n)=1

otherwise

where n is the reverse channel burst number relative to the start of thetransmission.

When the remote station transmits to the base, the base station expectsto receive each of the signals transmitted from the remotes at the samepower level. Thus, a gain control level is reported to the remotecontrol within the remote station from the DMT-SS demodulator. Thisautomatic gain control level (AGCL) is also transmitted from the DMT-SSmodulator to the up-converter so that the gain of the power amplifiercan be adjusted. In this manner, the base stations can assure that thesignal transmitted from the remote stations arrive at the base stationat the proper level.

The remote stations also have to perform synchronization. That is,although the remote stations are preprogrammed to operate within a TDDsystem, the specific information concerning the distinction between thetransmit and receive packets as well as the exact timing of the packettransfer still must be determined by the remote stations when a remotestation first comes on line. Subsequently, the remote stations mustacquire frequency synchronization for the DMT-SS signals so that theremotes are operating at the same frequency and phase as the basestation. For this reason, the DMT-SS demodulator generates a packetreference that is utilized by the synchronization circuitry to establishthe basic transmit/receive timing (i.e., the packet timing for the T/Rswitch). In addition, the packet timing is provided as a receive gate tothe demodulator and as a transmit gate to the modulator so that theremote station transmits and receives at the appropriate intervals.

Within the code-nulling network, measurements are taken on the waveformto determine the frequency error. The measured frequency error isprovided to the synchronization circuitry so that the remote station cancome into frequency and phase lock with the base station. Thissynchronization information is transmitted from the synchronizationcircuitry to the up-converter and the down-converter as a localoscillator reference and also as a digital-to-analog converter clock (orconversely, an analog-to-digital converter clock).

The code-nulling network also estimates the characteristics of themultitask channel (i.e., the frequency response of the multipathchannel). The channel estimates are provided to the spreading circuitryso that the preemphasis function can be performed to adaptively equalizethe multipath channel. Furthermore, the code-nulling network provides anestimate of the received power and the SINR to the remote controlcircuitry.

The resulting invention enables control over the power level of signalstransmitted by remote stations and base stations to minimizeinterference while insuring that signals reach their intendeddestination.

Although the preferred embodiments of the invention have been describedin detail above, it will be apparent to those of ordinary skill in theart that obvious modifications may be made to the invention withoutdeparting from its spirit or essence. Consequently, the precedingdescription should be taken as illustrative and not restrictive, and thescope of the invention should be determined in view of the followingclaims.

METHOD FOR NETWORK RETRODIRECTIVITY IN A DISCRETE MULTITONE SPREADSPECTRUM COMMUNICATIONS SYSTEM (2455-4383) BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention involves improvements to communications systems andmethods in a wireless discrete multitone spread spectrum communicationssystem.

2. Description of Related Art

Wireless communications systems, such as cellular and personalcommunications systems, operate over limited spectral bandwidths. Theymust make highly efficient use of the scarce bandwidth resource toprovide good service to a large population of users. Code DivisionMultiple Access (CDMA) protocol has been used by wireless communicationssystems to efficiently make use of limited bandwidths. The protocol usesa unique code to distinguish each user's data signal from other users'data signals. Knowledge of the unique code with which any specificinformation is transmitted, permits the separation and reconstruction ofeach user's message at the receiving end of the communication channel.

The personal wireless access network (PWAN) system described in thereferenced Alamouti, Stolarz, et al. patent application, uses a form ofthe CDMA protocol known as discrete multitone spread spectrum (DMT-SS)to provide efficient communications between a base station and aplurality of remote units. In this protocol, the user's data signal ismodulated by a set of weighted discrete frequencies or tones. Theweights are spreading codes that distribute the data signal over manydiscrete tones covering a broad range of frequencies. The weights arecomplex numbers with the real component acting to modulate the amplitudeof a tone while the complex component of the weight acts to modulate thephase of the same tone. Each tone in the weighted tone set bears thesame data signal. Plural users at the transmitting station can use thesame tone set to transmit their data, but each of the users sharing thetone set has a different set of spreading codes. The weighted tone setfor a particular user is transmitted to the receiving station where itis processed with despreading codes related to the user's spreadingcodes, to recover the user's data signal. For each of the spatiallyseparated antennas at the receiver, the received multitone signals aretransformed from time domain signals to frequency domain signals.Despreading weights are assigned to each frequency component of thesignals received by each antenna element. The values of the despreadingweights are combined with the received signals to obtain an optimizedapproximation of individual transmitted signals characterized by aparticular multitone set and transmitting location. The PWAN system hasa total of 2560 discrete tones (carriers) equally spaced in 8 MHZ ofavailable bandwidth in the range of 1850 to 1990 MHZ. The spacingbetween the tones is 3.125 kHz. The total set of tones are numberedconsecutively form 0 to 2559 starting from the lowest frequency tone.The tones are used to carry traffic messages and overhead messagesbetween the base station and the plurality of remote units. The traffictones are divided into 32 traffic partitions, with each traffic channelrequiring at least one traffic partition of 72 tones.

In addition, the PWAN system uses overhead tones to establishsynchronization and to pass control information between the base stationand the remote units. A Common Link Channel (CLC) is used by the base totransmit control information to the Remote Units. A Common AccessChannel (CAC) is used to transmit messages from the Remote Unit to theBase. There is one grouping of tones assigned to each channel. Theseoverhead channels are used in common by all of the remote units whenthey are exchanging control messages with the base station.

In the PWAN system, Time Division Duplexing (TDD) is used by the basestation and the remote unit to transmit data and control information inboth directions over the same multi-tone frequency channel. Transmissionfrom the base station to the remote unit is called forward transmissionand transmission from the remote unit to the base station is calledreverse transmission. The time between recurrent transmissions fromeither the remote unit or the base station is the TDD period. In everyTDD period, there are four consecutive transmission bursts in eachdirection. Data is transmitted in each burst using multiple tones. Thebase station and each remote unit must synchronize and conform to theTDD timing structure and both the base station and the remote unit mustsynchronize to a framing structure. All remote units and base stationsmust be synchronized so that all remote units transmit at the same timeand then all base stations transmit at the same time. When a remote unitinitially powers up, it acquires synchronization from the base stationso that it can exchange control and traffic messages within theprescribed TDD time format. The remote unit must also acquire frequencyand phase synchronization for the DMT-SS signals so that the remote isoperating at the same frequency and phase as the base station.

In the PWAN system, some of the tone frequencies are pilot tones used totransmit known symbols from the base station to the remote or from theremote station to the base, to enable synchronization of the stations.The referenced Hoole and the referenced Veintimilla patent applicationsdiscuss some of these features.

The PWAN system performs matrix operations involved in null-steering andcode-nulling procedures. The retrodirective weights calculated from thedespreading weights by the PWAN system are provided to transmitter pathfor use during data spreading, beam forming, and generating inverse fastFourier transforms (IFFTs).

In accordance with one aspect of the PWAN system, adaptive antennaarrays are used in conjunction with a beam forming algorithm to achievespatial diversity within each cell and implement SDMA. That is, signalsoutput by the antennas are directionally formed by selectivelyenergizing different antenna sensors with different signal gains so thatremote terminals in one portion of a cell are able to communicate withthe base station while other remote terminals in a different portion ofthe cell may communicate with the same base station, despite the factthat they are using the same tone set and code. It should be understoodthat in the fixed implementation of the current PWAN system, i.e., wherethe remote access terminals do not move substantially duringcommunication with the base station, usually staying within a cellduring communication, the beam forming algorithm used in the airlinkneed not account for mobile remote units leaving and entering the cell.In one advantageous embodiment, each cell is partitioned into foursectors where each sector transmits and receives over one of the foursub-band pairs.

As set forth above, the beam forming method of the PWAN system, like theuse of codes, should not be conceived as separate from the overalladaptive equalization method of the PWAN system. Rather, the method usedto selectively energize the antenna sensors (during transmission) orselectively weight the signals received on the different sensor elements(during reception) is subsumed into the overall method used to maximizeSINR. The relation of the beam forming method to the overallmaximization of SINR method will be described in greater detail below.

The use of spread-spectrum technology (particularly DMT-SS) anddirectional antennas within the preferred airlink of the PWAN systemallows for several error cancellation benefits, including effects thatare analogous to code nulling and null steering, by means of linearweighting in code and space.

Code-nulling is used to discriminate between non-orthogonal signalsemanating from adjacent cells. Again, the code-nulling method should beunderstood in the context of the maximization of SINR method of the PWANsystem. That is, the code-nulling method should be considered as theportion of the method that maximizes SINR with respect to the codedomain.

It should be understood that if signals generated within the same cellor beam all have orthogonal spreading codes, code-nulling is typicallynot necessary since the orthogonality is sufficient to ensure that thereis no cross modulation. However, as mentioned above, the spreading codesused within a particular cell may not be orthogonal, although they arepreferably linearly independent. Furthermore, the transceivers withinthe neighboring cells may employ spreading codes that have a randomcorrelation with the spreading codes used in the local cell.

By adjusting the spreading weights associated with each communicationschannel the base station is able to cross-correlate these signals on thesame tone set to subtract out interference due to “neighboring” signals.In one embodiment, the base station has the spreading codes used tospread different signals assigned to the same tone set, so that thisinformation can be used to initially calculate the appropriate weightsfor nulling out interference from other codes.

As discussed above, when the spreading codes used to spread distinctdata signals are orthogonal, the spread data can be precisely recoveredduring despreading. However, when the spreading codes are not orthogonal(as is the case with spreading codes that are used in neighboringcells), cross modulation may result so that the data signals are notable to be precisely distinguished by simple despreading (i.e.,despreading without code-nulling).

In order to compensate for this phenomena, code-nulling weights aremultiplied by the received signal vector. By nulling out the crossmodulation present in the received signal, the appropriate values of thedata bits are output by the receiver. As long as the complex spreadingweights are linearly independent the exact symbol values can bediscriminated by this method. It will be appreciated that thecode-nulling procedure above is inherently implemented during derivationof the overall weights that maximize the SINR.

In addition to code-nulling, the exemplary directional antenna shown inFIGS. 11 and 12 forms signals including null regions (i.e., regionswhere the antenna attenuates incoming signals or where there is a verylow antenna gain). These null regions can be formed in a pattern so thatthe nulls are directed towards known interferers (e.g., interferingsignal sources or interfering multi-path reflectors). In this manner,interfering signals are de-emphasized in the spatial domain. As will bediscussed in greater detail below, the use of null-steering inconjunction with code-nulling is highly advantageous.

In accordance with one aspect of the PWAN system, significant processingtime and sophistication can be saved since significant similarity existsbetween the methods for performing null-steering and code-nulling.Specifically, the mathematical formalism used to achieve null-steeringis analogous to the formalism used to achieve code-nulling. According tothis analogy, just as the tones in a tone set are multiplied by complexweights to alter the amplitude and phase of the tones, so are the gainand relative phase of signals output and received by the antennaelements altered by a set of multiplicative weights. This multiplicationby complex weights can be expressed in a matrix form for both codenulling—a spectral concept—and null steering—a spatial concept. Thus,the calculations performed in the spectral code domain correspondformally to the calculations performed in the spatial domain.Consequently, null steering can be performed in a system usingcode-nulling simply by adding an extra dimension to the matrices usedfor calculating the complex weights and multiplying the signals by theseweights.

What is needed in wireless communications systems composed of multiplecells, is the ability to optimize the entire network for inter-cellinterference.

SUMMARY OF THE INVENTION

The entire network of multiple base station cells achieves anequilibrium state where every communications link in every cell reachesan optimized signal interference level. Each base station and eachremote unit in a cell adapts its despreading weights to minimizeinterference from other cells. Then each base station and each remotestation in a cell adapts its spreading weights to minimize sendinginterfering signals to other cells. This creates a network-wide couplingof the cells that results in optimizing the entire network forinter-cell interference.

In one aspect of the invention, optimum weights are calculated based onall of the signals received at the base station. Since the set of tonefrequencies on the receive path is the same as the set of tonefrequencies on the transmit path, the despreading weights used toreceive can be used to compute the spreading weights for transmission.This is the principle of retrodirectivity. In addition, the value of thedespreading weights are adaptively computed, adjusted to minimizereceive sensitivity to interfering signals. The spreading weightsderived from the despreading weights are also adaptive, their valuesbeing adjusted to diminish the strength of signals transmitted back inthe direction of the interfering signal source. Null steering and codenulling are used to adjust the despreading weights and the spreadingweights to adaptively minimize the exchange of interfering signals. Whenadaptive retrodirectivity is used to determine the set of weights forboth reception and transmission in each cell of the network,network-wide adaptive retrodirectivity can be accomplished. The basestations and remote stations in each cell use null-steering and codenulling to diminish their interference with stations in other cells. Theretrodirective formation of spreading weights from despreading weightsin each station propagates channel optimization across cell boundaries.This coupling of cells using the principle of retrodirectivity optimizesthe channel characteristics throughout the entire system.

Currently, the invention has advantageous applications in the field ofwireless communications such as cellular communications or personalcommunications, where bandwidth is scarce compared to the number of theusers and their needs,. Such applications may be effected in mobile,fixed or minimally mobile systems. However, the invention may beadvantageously applied to other, non-wireless, communications systems aswell.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 102 is a network diagram of two cells engaging in a first stage ofretrodirective coupling, where the base station B1 in cell 1 detects thepresence of interfering signals from the remote station R2 in theneighboring cell 2. Base station B1 adjusts its transmissions in thedirection of remote station R2 to diminish their signal strength.

FIG. 103 is a network diagram of the two cells of FIG. 102 in a secondstage of retrodirective coupling, where the base station B2 in thesecond cell 2 detects the presence of interfering signals from theremote station R1′ in the first cell 1. Base station B2 adjusts itstransmissions in the direction of remote station R1′ to diminish theirsignal strength.

FIG. 104 is a network diagram of the four cells similar to FIGS. 1A and1B, showing propagation of channel optimization across cell boundariesto optimize the channel characteristics throughout the entire system.

FIG. 105 is a more detailed block diagram of base station B1 and remotestation R1′ in cell 1 and remote station R2 in cell 2, where remotestation R2 is sending interfering signals to the base station B1.

FIG. 106 is a detailed block diagram similar to FIG. 105, showing thebase station B1 sending diminished strength signals across the cellboundary, in the direction of the interfering remote station R2.

DISCUSSION OF THE PREFERRED EMBODIMENT

FIG. 102 is a network diagram of two cells 1 and 2, in a PWANcommunications system. Base station B1 communicates with remote stationsR1 and R1′ using the DMT-SS protocol. The notation (B1->R1′) indicatesthe path from base station B1 to the remote station R1′, for example.The notation (R1′->B1) indicates the path from remote station R1′ backto the base station B1. The notation (R2->B1) indicates the path fromremote station R2 in the neighboring cell 2 to the base station B1. Basestation B1 in cell 1 detects the presence of interfering signals fromthe remote station R2 in the neighboring cell 2. In accordance with theinvention, base station B1 adjusts its transmissions in the direction ofremote station R2 to diminish their signal strength.

FIG. 105 is a more detailed block diagram of base station B1 and remotestation R1′ in cell 1 and remote station R2 in cell 2, where remotestation R2 is sending interfering signals to the base station B1. Remotestation R1′ in the same cell as base station B1, sends data tones andpilot tones to base station B1 using the DMT-SS protocol. Remote stationR2 in the neighboring cell 2 sends an interfering signal to base stationB1, also using the DMT-SS protocol. The base station B1 calculatesoptimum weights based on all of the signals received at the base stationusing the adaptive processor. Since the set of tone frequencies on thereceive path is the same as the set of tone frequencies on the transmitpath, the despreading weights used to receive can be used to compute thespreading weights for transmission, using the principle ofretrodirectivity. The adaptive processor computes the value of thedespreading weights, adjusted to minimize receive sensitivity tointerfering signals from remote station R2.

FIG. 106 is a detailed block diagram similar to FIG. 105, showing thebase station B1 sending diminished strength signals across the cellboundary, in the direction of the interfering remote station R2. Thespreading weights derived from the despreading weights are alsoadaptive, their values being adjusted to diminish the strength ofsignals transmitted back in the direction of the interfering signalsource, R2. Null steering and code nulling are used to adjust thedespreading weights and the spreading weights to adaptively minimize theexchange of interfering signals.

FIG. 102 shows base station B2 communicating with remote stations R2 andR2′ using the DMT-SS protocol. FIG. 103 is a network diagram of the twocells of FIG. 102 in a second stage of retrodirective coupling, wherethe base station B2 in the second cell 2 detects the presence ofinterfering signals from the remote station R1′ in the first cell 1.Base station B2 adjusts its transmissions in the direction of remotestation R1′ to diminish their signal strength. When adaptiveretrodirectivity is used to determine the set of weights for bothreception and transmission in each cell of the network, network-wideadaptive retrodirectivity can be accomplished. The base stations andremote stations in each cell use null-steering and code nulling todiminish their interference with stations in other cells. Theretrodirective formation of spreading weights from despreading weightsin each station propagates channel optimization across cell boundaries.

FIG. 104 is a network diagram of the four cells 1, 2, 3, and 4, similarto FIGS. 1A and 1B, showing propagation of channel optimization acrosscell boundaries to optimize the channel characteristics throughout theentire system.

FIG. 102 also shows how the remote station R2 in cell 2 responds to thepresence of interference signals it detects from the base station incell 1, to optimize the multiple cell network for inter-cellinterference. As was discussed above, base station B1 is receiving afirst spread signal comprising a first data signal spread over aplurality of discrete tones received over a first path (R1′->B1) fromremote station R1′ located in cell 1. The first signal further includesan interfering signal spread over the plurality of discrete tonesreceived over an interference path (R2->B1) from remote station R2located in cell 2. Base station B1 is adaptively despreading the signalreceived by using first despreading codes that are based on thecharacteristics of the received spread signal over the first path(R1′->B1) and over the interference path (R2->B1). The base station B1then is spreading a second data signal with first spreading codesderived from the despreading codes based on the retrodirectivity of thefirst path (R1′->B1) and of the interference path (R2->B1). The firstspreading codes are distributing the second data signal over a pluralityof discrete tones, forming a second spread signal that is selectivelydiminished in the interfering path (B1->R2) to the second remote stationR2. Then base station B1 continues by transmitting the second spreadsignal over the first path (B1->R1′) to the first remote station R1′ andtransmitting the second signal selectively diminished over theinterference path (B1->R2) to the second remote station R2.

Then the remote station R2 in cell 2 receives the selectively diminishedsecond spread signal. Remote station R2 then continues by adaptivelydespreading the selectively diminished second signal it received byusing second despreading codes that are based on the characteristics ofthe received second signal over the interference path (B1 ->R2). Theremote station R2 then continues by spreading a third data signal withsecond spreading codes derived from the second despreading codes basedon a retrodirectivity of the interference path (B1->R2). The secondspreading codes distribute the third data signal over a plurality ofdiscrete tones, forming a third spread signal that is selectivelydiminished in the interfering path (R2->B1) to the first base stationB1. The remote station R2 then continues by transmitting the selectivelydiminished third spread signal over the interference path (R2->B1) tothe first base station B1. In this manner the remote station R2 in thesecond cell 2 modifies its despreading and spreading weights to minimizethe exchange of interference signals across the cell boundary to basestation B1.

Although the preferred embodiments of the invention have been describedin detail above, it will be apparent to those of ordinary skill in theart that obvious modifications may be made to the invention withoutdeparting from its spirit or essence. Consequently, the precedingdescription should be taken as illustrative and not restrictive, and thescope of the invention should be determined in view of the followingclaims.

What is claimed is:
 1. A highly bandwidth-efficient communicationsmethod, comprising the steps of: receiving at a base station a spreadsignal comprising an incoming data traffic signal spread over aplurality of discrete traffic frequencies and an incoming messagesegment signal spread over a plurality of link control frequencies;adaptively despreading the signals received at the base station by usingdespreading weights; detecting a priority interrupt flag value in saidmessage segment signal; resetting a message segment buffer in said basestation and storing a message segment therein, if said priorityinterrupt flag has a first value; concatenating said message segmentwith a previously received message segment, if said priority interruptflag has a second value.
 2. The highly bandwidth-efficientcommunications method of claim 1, wherein said base station is part of awireless discrete multitone spread spectrum communications system. 3.The highly bandwidth-efficient communications method of claim 1, whereinsaid message segment is a system management message segment.
 4. Thehighly bandwidth-efficient communications method of claim 1, whereinsaid first value of said priority interrupt flag corresponds to a timecritical message segment.
 5. The highly bandwidth-efficientcommunications method of claim 1, wherein said second value of saidpriority interrupt flag corresponds to a message segment that is not afirst segment.
 6. A highly bandwidth-efficient communications method,comprising the steps of: receiving at a base station a first spreadsignal comprising an incoming data traffic signal having a data portionspread over a plurality of discrete traffic frequencies; receiving atsaid base station a second spread signal comprising an incoming messagesegment signal having a message segment portion and a priority interruptflag portion spread over a plurality of link control frequencies;adaptively despreading said first spread signal received at the basestation by using despreading weights, recovering said data portion;adaptively despreading said second spread signal received at the basestation by using despreading weights, recovering said message segmentportion and said priority interrupt flag portion; resetting a messagesegment buffer in said base station and storing said message segmentportion therein, if said priority interrupt flag has a first value;concatenating said message segment portion with a previously receivedmessage segment, if said priority interrupt flag has a second value. 7.The highly bandwidth-efficient communications method of claim 6, whereinsaid base station is part of a wireless discrete multitone spreadspectrum communications system.
 8. The highly bandwidth-efficientcommunications method of claim 6, wherein said message segment is asystem management message segment.
 9. The highly bandwidth-efficientcommunications method of claim 6, wherein said first value of saidpriority interrupt flag corresponds to a time critical message segment.10. The highly bandwidth-efficient communications method of claim 6,wherein said second value of said priority interrupt flag corresponds toa message segment that is not a first segment.
 11. A highlybandwidth-efficient communications method, comprising the steps of:transmitting from a base station a transmitted spread signal comprisingan outgoing data traffic signal spread over a plurality of discretetraffic frequencies and an outgoing message segment signal spread over aplurality of link control frequencies; said outgoing message segmentsignal being part of a low priority message having a second outgoingmessage segment signal to be transmitted; receiving at said base stationa spread signal comprising an incoming data traffic signal spread over aplurality of discrete traffic frequencies and an incoming messagesegment signal spread over a plurality of link control frequencies;adaptively despreading the signals received at the base station by usingdespreading weights; detecting a priority interrupt flag value in saidmessage segment signal; interrupting transmission of said secondoutgoing message segment signal, resetting a message segment buffer insaid base station, and storing said incoming message segment signaltherein, if said priority interrupt flag has a first value;concatenating said incoming message segment signal with a previouslyreceived message segment, if said priority interrupt flag has a secondvalue.
 12. The highly bandwidth-efficient communications method of claim11, wherein said message segment signals are system management messagesegment signals.
 13. The highly bandwidth-efficient communicationsmethod of claim 11, wherein said first value of said priority interruptflag corresponds to a time critical message segment signal.
 14. Thehighly bandwidth-efficient communications method of claim 11, whereinsaid second value of said priority interrupt flag corresponds to amessage segment signal that is not a first segment.
 15. A highlybandwidth-efficient communications method, comprising the steps of:transmitting from a base station a transmitted spread signal comprisingan outgoing data traffic signal spread over a plurality of discretetraffic frequencies and an outgoing message segment signal spread over aplurality of link control frequencies; said outgoing message segmentsignal being part of a low priority message having a second outgoingmessage segment signal to be transmitted; receiving at a base station afirst spread signal comprising an incoming data traffic signal having adata portion spread over a plurality of discrete traffic frequencies;receiving at said base station a second spread signal comprising anincoming message segment signal having a message segment portion and apriority interrupt flag portion spread over a plurality of link controlfrequencies; adaptively despreading said first spread signal received atthe base station by using despreading weights, recovering said dataportion; adaptively despreading said second spread signal received atthe base station by using despreading weights, recovering said messagesegment portion and said priority interrupt flag portion; interruptingtransmission of said second outgoing message segment signal, resetting amessage segment buffer in said base station and storing said messagesegment portion therein, if said priority interrupt flag has a firstvalue; concatenating said message segment portion with a previouslyreceived message segment, if said priority interrupt flag has a secondvalue.
 16. The highly bandwidth-efficient communications method of claim15, wherein said station is part of a wireless discrete multitone spreadspectrum communications system.
 17. The highly bandwidth-efficientcommunications method of claim 15, wherein said message segment is asystem management message segment.
 18. The highly bandwidth-efficientcommunications method of claim 15, wherein said first value of saidpriority interrupt flag corresponds to a time critical message segment.19. The highly bandwidth-efficient communications method of claim 15,wherein said second value of said priority interrupt flag corresponds toa message segment that is not a first segment.
 20. A highlybandwidth-efficient communications method, comprising the steps of:receiving at a station a spread signal comprising an incoming datatraffic signal spread over a plurality of discrete traffic frequenciesand an incoming message segment signal spread over a plurality of linkcontrol frequencies; adaptively despreading the signals received at thestation by using despreading weights; detecting a priority interruptflag value in said message segment signal; resetting a message segmentbuffer in said station and storing a message segment therein, if saidpriority interrupt flag has a first value; concatenating said messagesegment with a previously received message segment, if said priorityinterrupt flag has a second value.
 21. The highly bandwidth-efficientcommunications method of claim 20, wherein said station is part of awireless discrete multitone spread spectrum communications system. 22.The highly bandwidth-efficient communications method of claim 20,wherein said message segment is a system management message segment. 23.The highly bandwidth-efficient communications method of claim 20,wherein said first value of said priority interrupt flag corresponds toa time critical message segment.
 24. The highly bandwidth-efficientcommunications method of claim 20, wherein said second value of saidpriority interrupt flag corresponds to a message segment that is not afirst segment.
 25. The highly bandwidth-efficient communications methodof claim 20, wherein said station is a base station in a wirelessdiscrete multitone spread spectrum communications system.
 26. The highlybandwidth-efficient communications method of claim 20, wherein saidstation is a remote station in a wireless discrete multitone spreadspectrum communications system.
 27. A highly bandwidth-efficientcommunications method, comprising the steps of: receiving at a station afirst spread signal comprising an incoming data traffic signal having adata portion spread over a plurality of discrete traffic frequencies;receiving at said station a second spread signal comprising an incomingmessage segment signal having a message segment portion and a priorityinterrupt flag portion spread over a plurality of link controlfrequencies; adaptively despreading said first spread signal received atthe station by using despreading weights, recovering said data portion;adaptively despreading said second spread signal received at the stationby using despreading weights, recovering said message segment portionand said priority interrupt flag portion; resetting a message segmentbuffer in said station and storing said message segment portion therein,if said priority interrupt flag has a first value; concatenating saidmessage segment portion with a previously received message segment, ifsaid priority interrupt flag has a second value.
 28. The highlybandwidth-efficient communications method of claim 27, wherein saidstation is a base station in a wireless discrete multitone spreadspectrum communications system.
 29. The highly bandwidth-efficientcommunications method of claim 27, wherein said station is a remotestation in a wireless discrete multitone spread spectrum communicationssystem.
 30. A highly bandwidth-efficient communications system,comprising: means for receiving at a base station a spread signalcomprising an incoming data traffic signal spread over a plurality ofdiscrete traffic frequencies and an incoming message segment signalspread over a plurality of link control frequencies; means foradaptively despreading the signals received at the base station by usingdespreading weights; means for detecting a priority interrupt flag valuein said message segment signal; means for resetting a message segmentbuffer in said base station and storing a message segment therein, ifsaid priority interrupt flag has a first value; means for concatenatingsaid message segment with a previously received message segment, if saidpriority interrupt flag has a second value.
 31. The highlybandwidth-efficient communications system of claim 30, wherein said basestation is part of a wireless discrete multitone spread spectrumcommunications system.
 32. The highly bandwidth-efficient communicationssystem of claim 30, wherein said message segment is a system managementmessage segment.
 33. The highly bandwidth-efficient communicationssystem of claim 30, wherein said first value of said priority interruptflag corresponds to a time critical message segment.
 34. The highlybandwidth-efficient communications system of claim 30, wherein saidsecond value of said priority interrupt flag corresponds to a messagesegment that is not a first segment.
 35. A highly bandwidth-efficientcommunications system, comprising: means for receiving at a station afirst spread signal comprising an incoming data traffic signal having adata portion spread over a plurality of discrete traffic frequencies;means for receiving at said station a second spread signal comprising anincoming message segment signal having a message segment portion and apriority interrupt flag portion spread over a plurality of link controlfrequencies; means for adaptively despreading said first spread signalreceived at the station by using despreading weights, recovering saiddata portion; means for adaptively despreading said second spread signalreceived at the station by using despreading weights, recovering saidmessage segment portion and said priority interrupt flag portion; meansfor resetting a message segment buffer in said station and storing saidmessage segment portion therein, if said priority interrupt flag has afirst value; means for concatenating said message segment portion with apreviously received message segment, if said priority interrupt flag hasa second value.
 36. The highly bandwidth-efficient communications systemof claim 35, wherein said station is a base station in a wirelessdiscrete multitone spread spectrum communications system.
 37. The highlybandwidth-efficient communications system of claim 35, wherein saidstation is a remote station in a wireless discrete multitone spreadspectrum communications system.
 38. A highly bandwidth-efficientcommunications method, comprising the steps of: receiving at a station aspread signal comprising an incoming message segment signal spread overa plurality of frequencies; adaptively despreading the signals receivedat the station by using despreading weights; detecting a priorityinterrupt flag value in said message segment signal; resetting a messagesegment buffer in said station and storing a message segment therein, ifsaid priority interrupt flag has a first value; and concatenating saidmessage segment with a previously received message segment, if saidpriority interrupt flag has a second value.
 39. A highlybandwidth-efficient communications method, comprising the steps of:receiving at said station a spread signal comprising an incoming messagesegment signal having a message segment portion and a priority interruptflag portion spread over a plurality of frequencies; adaptivelydespreading said spread signal received at the station by usingdespreading weights, recovering said message segment portion and saidpriority interrupt flag portion; resetting a message segment buffer insaid station and storing said message segment portion therein, if saidpriority interrupt flag has a first value; concatenating said messagesegment portion with a previously received message segment, if saidpriority interrupt flag has a second value.
 40. A highlybandwidth-efficient communications method, comprising the steps of:transmitting from a station a transmitted spread signal comprising anoutgoing message segment signal spread over a plurality of frequencies;said outgoing message segment signal being part of a low prioritymessage having a second outgoing message segment signal to betransmitted; receiving at said station a spread signal comprising anincoming message segment signal spread over a plurality of frequencies;adaptively despreading the signals received at the station by usingdespreading weights; detecting a priority interrupt flag value in saidmessage segment signal; interrupting transmission of said secondoutgoing message segment signal, resetting a message segment buffer insaid station, and storing said incoming message segment signal therein,if said priority interrupt flag has a first value; concatenating saidincoming message segment signal with a previously received messagesegment, if said priority interrupt flag has a second value.
 41. Ahighly bandwidth-efficient communications method, comprising the stepsof: transmitting from a station a transmitted spread signal comprisingan outgoing message segment signal spread over a plurality offrequencies; said outgoing message segment signal being part of a lowpriority message having a second outgoing message segment signal to betransmitted; receiving at said station a spread signal comprising anincoming message segment signal having a message segment portion and apriority interrupt flag portion spread over a plurality of frequencies;adaptively despreading said received spread signal received at thestation by using despreading weights, recovering said message segmentportion and said priority interrupt flag portion; interruptingtransmission of said second outgoing message segment signal, resetting amessage segment buffer in said station and storing said message segmentportion therein, if said priority interrupt flag has a first value; andconcatenating said message segment portion with a previously receivedmessage segment, if said priority interrupt flag has a second value. 42.A highly bandwidth-efficient communications method, comprising the stepsof: receiving at a station a spread signal comprising an incomingmessage segment signal spread over a plurality of link controlfrequencies; adaptively despreading the signals received at the stationby using despreading weights; detecting a priority interrupt flag valuein said message segment signal; storing a message segment if saidpriority interrupt flag has a first value; concatenating said messagesegment with a previously received message segment, if said priorityinterrupt flag has a second value.
 43. A highly bandwidth-efficientcommunications method, comprising the steps of: receiving at saidstation a spread signal comprising an incoming message segment signalhaving a message segment portion and a priority interrupt flag portionspread over a plurality of frequencies; adaptively despreading saidspread signal received at the station by using despreading weights,recovering said message segment portion and said priority interrupt flagportion; storing said message segment portion if said priority interruptflag has a first value; concatenating said message segment portion witha previously received message segment, if said priority interrupt flaghas a second value.
 44. A highly bandwidth-efficient communicationssystem, comprising: means for receiving at a station a spread signalcomprising an incoming message segment signal spread over a plurality offrequencies; means for adaptively despreading the signals received atthe station by using despreading weights; means for detecting a priorityinterrupt flag value in said message segment signal; means for storing amessage segment if said priority interrupt flag has a first value; andmeans for concatenating said message segment with a previously receivedmessage segment, if said priority interrupt flag has a second value. 45.A highly bandwidth-efficient communications system, comprising: meansfor receiving at said station a spread signal comprising an incomingmessage segment signal having a message segment portion and a priorityinterrupt flag portion spread over a plurality of frequencies; means foradaptively despreading said spread signal received at the station byusing despreading weights, recovering said message segment portion andsaid priority interrupt flag portion; means for resetting a messagesegment buffer in said station and storing said message segment portiontherein, if said priority interrupt flag has a first value; and meansfor concatenating said message segment portion with a previouslyreceived message segment, if said priority interrupt flag has a secondvalue.